| United States Patent |
6,515,740
|
|
Bamji
, et al.
|
February 4, 2003
|
Methods for CMOS-compatible three-dimensional image sensing using quantum
efficiency modulation
Abstract
A preferably CMOS-implementable method measures distance and/or brightness
by illuminating a target with emitted optical energy having a modulated
periodic waveform whose high frequency component may be idealized as
S.sub.1 =cos(.omega..multidot.t). A fraction of the emitted optical energy
is reflected by a target and detected with at least one in a plurality of
semiconductor photodetectors. Photodetector quantum efficiency is
modulated to process detected signals to yield data proportional to the
distance z separating the target and photodetector. Detection includes
measuring phase change between the emitted optical energy and the
reflected fraction thereof. Quantum efficiency can be modulated with fixed
or variable phase methods and may be enhanced using enhanced photocharge
collection, differential modulation, and spatial and temporal
multiplexing. System power requirements may be reduced with inductors that
resonate with photodetector capacitance at the operating frequency. The
method can be implemented with on-chip photodetectors, associated
electronics, and processing.
| Inventors:
|
Bamji; Cyrus (Fremont, CA);
Charbon; Edoardo (Berkeley, CA)
|
| Assignee:
|
Canesta, Inc. (San Jose, CA)
|
| Appl. No.:
|
020393 |
| Filed:
|
December 11, 2001 |
| Current U.S. Class: |
356/141.1; 356/4.01; 356/5.1 |
| Intern'l Class: |
G01C 003/08; G01C 001/00; G01B 011/26 |
| Field of Search: |
356/141.1,5.1,4.01
|
References Cited [Referenced By]
U.S. Patent Documents
Primary Examiner: Buczinski; Stephen C.
Attorney, Agent or Firm: Dorsey & Whitney LLP
Parent Case Text
RELATION TO PREVIOUSLY FILED APPLICATIONS
Priority is claimed from applicants' co-pending U.S. provisional patent
application serial No. 60/254,873 filed on Dec. 11, 2000 entitled "CMOS 3D
Multi-Pixel Sensor Using Photodiode Quantum Efficiency Modulation" and
60/247,158, filed Nov. 9, 2000. Applicants incorporate said application
herein by reference. Applicants also refer to and incorporates by
reference herein co-pending U.S. utility application Ser. number
09/876,373 filed Jun. 6, 2001 entitled "CMOS-Compatible Three-Dimensional
Image Sensing Using Reduced Peak Energy".
Claims
What is claimed is:
1. A method to determine distance z between at least one photodetector, and
a target, the method comprising the following steps:
(a) illuminating said target with optical energy that has a modulated
periodic waveform that includes a high frequency component S.sub.1
(.omega..multidot.t);
(b) detecting with said photodetector a fraction of said optical energy
reflected from said target; and
(c) modulating quantum efficiency of said photodetector to process signals
detected at step (b) to yield data proportional to said distance z.
2. The method of claim 1, further including a plurality of photodetectors
fabricated on an integrated circuit chip;
wherein said integrated circuit chip includes circuitry that carries out
step (b) and step (c).
3. The method of claim 1, wherein said plurality includes at least one of
(i) photodiode detectors, (ii) MOS devices with a bias gate, and (iii) MOS
devices with a photogate.
4. The method of claim 1, wherein detecting at step (b) includes measuring
a change in phase between optical energy emitted at step (a) and a signal
detected at step (b).
5. The method of claim 4, wherein step (c) includes using a variable phase
delay that is coupled to a source of said modulated periodic waveform,
operating in a closed-loop, such that phase delay of said variable phase
delay indicates phase delay of a signal detected at step (b).
6. The method of claim 4, wherein step (c) includes use of at least one
fixed phase delay.
7. The method of claim 4, wherein said change of phase is proportional to
said distance z.
8. The method of claim 1, wherein step (c) includes varying reverse bias of
said photodetectors.
9. The method of claim 1, wherein said photodetectors include photogate
detectors, and step (c) includes varying gate potential of said photogate
detectors.
10. The method of claim 1, wherein:
detecting at step (b) includes measuring a change in phase between optical
energy emitted at step (a) and a signal detected at step (b); further
including:
defining banks of said photodetectors; and
enhancing efficiency of said quantum efficiency modulation by modulating
banks of said photodetectors with different phases.
11. The method of claim 1, wherein said photodetectors are formed on a
semiconductor substrate; and
step (c) includes creating an electrical current in said substrate to
promote collection of photocharges released within said substrate by
reflected said optical energy;
wherein quantum efficiency modulation is enhanced.
12. The method of claim 1, wherein said photodetectors are formed on a
semiconductor substrate including an epitaxial region; and
step (c) includes using a substrate whose said epitaxial region has at
least one characteristic selected from (i) said epitaxial region comprises
a plurality of layers each having a different doping concentration,
wherein an uppermost one of said layers is less highly doped than a lower
one of said layers, (ii) said epitaxial region defines a layer in which
there is a dopant gradient such that doping concentration is greater at a
lower portion of said region than at an upper portion thereof.
13. The method of claim 1, further including coupling an inductor so as to
detune at least some capacitance coupled to a voltage node of said
detector controlling quantum efficiency modulation thereof;
wherein power dissipation of said capacitance is reduced.
14. The method of claim 1, further including:
defining at least a first bank of said photodetectors and a second bank of
said photodetectors, each said bank being quantum efficiency modulated
with a constant phase;
defining at least one pixel comprising a said photodetector from said first
bank and from said second bank;
wherein step (c) includes processing an output from one said photodetector
for use by more than one said pixel.
15. The method of claim 1, wherein:
distance z in determined over multiple time frames; and
claim (c) further includes:
on a per frame basis, quantum efficiency modulating said photodetector with
at least a first phase shift, and acquiring information from said
photodetector during said first phase shift; and
wherein information acquired from said photodetector during said first
phase shift is used in at least two said time frames.
16. The method of claim 1, further including:
digitizing an analog output from each said photodetector.
17. The method of claim 1, wherein said frequency .omega. is at least 100
MHz.
18. The method of claim 1, further including providing an integrated
circuit that includes electronic circuitry that carries out at least one
of step (b) and step (c).
19. The method of claim 1, wherein step (a) includes illuminating said
target with optical energy having wavelength of about 850 nm.
20. A method to determine amplitude of a fraction of emitted optical energy
that is reflected from a target, the method comprising the following
steps:
(a) illuminating said target with optical energy that has a modulated
periodic waveform that includes a high frequency component S.sub.1
(.omega..multidot.t);
(b) providing at least one photodetector to detect said fraction of optical
energy reflected by said target;
(c) detecting with said photodetector said fraction of said optical energy
reflected from said target; and
(d) modulating quantum efficiency of said photodetector to process signals
detected at step (c) to yield data proportional to amplitude.
21. The method of claim 20, wherein said frequency .omega. is at least 100
Hz.
22. The method of claim 20, wherein step (a) includes illuminating said
target with optical energy having wavelength of about 850 nm.
Description
FIELD OF THE INVENTION
The invention relates generally to range finder type image sensors,
especially range finder image sensors that are implementable on a single
integrated circuit using CMOS fabrication, and more particularly to
reducing power consumption of systems utilizing such sensors.
BACKGROUND OF THE INVENTION
Electronic circuits that provide a measure of distance from the circuit to
an object are known in the art, and may be exemplified by system 10 FIG.
1. In the generalized system of FIG. 1, imaging circuitry within system 10
is used to approximate the distance (e.g., Z1, Z2, Z3) to an object 20,
the top portion of which is shown more distant from system 10 than is the
bottom portion. Typically system 10 will include a light source 30 whose
light output is focused by a lens 40 and directed toward the object to be
imaged, here object 20. Other prior art systems do not provide an active
light source 30 and instead rely upon and indeed require ambient light
reflected by the object of interest.
Various fractions of the light from source 30 may be reflected by surface
portions of object 20, and is focused by a lens 50. This return light
falls upon various detector devices 60, e.g., photodiodes or the like, in
an array on an integrated circuit (IC) 70. Devices 60 produce a rendering
of the luminosity of an object (e.g., 10) in the scene from which distance
data is to be inferred. In some applications devices 60 might be charge
coupled devices (CCDs) or even arrays of CMOS devices.
CCDs typically are configured in a so-called bucket-brigade whereby
light-detected charge by a first CCD is serial-coupled to an adjacent CCD,
whose output in turn is coupled to a third CCD, and so on. This
bucket-brigade configuration generally precludes fabricating processing
circuitry on the same IC containing the CCD array. Further, CCDs provide a
serial readout as opposed to a random readout. For example, if a CCD range
finder system were used in a digital zoom lens application, even though
most of the relevant data would be provided by a few of the CCDs in the
array, it would nonetheless be necessary to readout the entire array to
gain access to the relevant data, a time consuming process. In still and
some motion photography applications, CCD-based systems might still find
utility.
As noted, the upper portion of object 20 is intentionally shown more
distant that the lower portion, which is to say distance Z3>Z2>Z1.
In a range finder autofocus camera environment, one might try to have
devices 60 approximate average distance from the camera (e.g., from Z=0)
to object 10 by examining relative luminosity data obtained from the
object. In some applications, e.g., range finding binoculars, the field of
view is sufficiently small such that all objects in focus will be at
substantially the same distance. But in general, luminosity-based systems
do not work well. For example, in FIG. 1, the upper portion of object 20
is shown darker than the lower portion, and presumably is more distant
than the lower portion. But in the real world, the more distant portion of
an object could instead be shinier or brighter (e.g., reflect more optical
energy) than a closer but darker portion of an object. In a complicated
scene, it can be very difficult to approximate the focal distance to an
object or subject standing against a background using change in luminosity
to distinguish the subject from the background. In such various
applications, circuits 80, 90, 100 within system 10 in FIG. 1 would assist
in this signal processing. As noted, if IC 70 includes CCDs 60, other
processing circuitry such as 80, 90, 100 are formed off-chip.
Unfortunately, reflected luminosity data does not provide a truly accurate
rendering of distance because the reflectivity of the object is unknown.
Thus, a distant object surface with a shiny surface may reflect as much
light (perhaps more) than a closer object surface with a dull finish.
Other focusing systems are known in the art. Infrared (IR) autofocus
systems for use in cameras or binoculars produce a single distance value
that is an average or a minimum distance to all targets within the field
of view. Other camera autofocus systems often require mechanical focusing
of the lens onto the subject to determine distance. At best these prior
art focus systems can focus a lens onto a single object in a field of
view, but cannot simultaneously measure distance for all objects in the
field of view.
In general, a reproduction or approximation of original luminosity values
in a scene permits the human visual system to understand what objects were
present in the scene and to estimate their relative locations
stereoscopically. For non-stereoscopic images such as those rendered on an
ordinary television screen, the human brain assesses apparent size,
distance and shape of objects using past experience. Specialized computer
programs can approximate object distance under special conditions.
Stereoscopic images allow a human observer to more accurately judge the
distance of an object. However it is challenging for a computer program to
judge object distance from a stereoscopic image. Errors are often present,
and the required signal processing requires specialized hardware and
computation. Stereoscopic images are at best an indirect way to produce a
three-dimensional image suitable for direct computer use.
Many applications require directly obtaining a three-dimensional rendering
of a scene. But in practice it is difficult to accurately extract distance
and velocity data along a viewing axis from luminosity measurements.
Nonetheless many applications require accurate distance and velocity
tracking, for example an assembly line welding robot that must determine
the precise distance and speed of the object to be welded. The necessary
distance measurements may be erroneous due to varying lighting conditions
and other shortcomings noted above. Such applications would benefit from a
system that could directly capture three-dimensional imagery.
Although specialized three dimensional imaging systems exist in the nuclear
magnetic resonance and scanning laser tomography fields, such systems
require substantial equipment expenditures. Further, these systems are
obtrusive, and are dedicated to specific tasks, e.g., imaging internal
body organs.
In other applications, scanning laser range finding systems raster scan an
image by using mirrors to deflect a laser beam in the x-axis and perhaps
the y-axis plane. The angle of defection of each mirror is used to
determine the coordinate of an image pixel being sampled. Such systems
require precision detection of the angle of each mirror to determine which
pixel is currently being sampled. Understandably having to provide
precision moving mechanical parts add bulk, complexity, and cost to such
range finding system. Further, because these systems sample each pixel
sequentially, the number of complete image frames that can be sampled per
unit time is limited. It is understood that the term "pixel" can refer to
an output result produced from one or more detectors in an array of
detectors.
In summation, there is a need for a method and system that can produce
direct three-dimensional imaging, preferably using circuitry that can be
fabricated on a single IC using CMOS fabrication techniques, and requiring
few discrete components and no moving components. Optionally, the system
should be able to output data from the detectors in a non-sequential or
random fashion. Very preferably, such system should require relatively low
peak light emitting power such that inexpensive light emitters may be
employed, yet the system should provide good sensitivity.
The present invention provides such a method.
SUMMARY OF THE PRESENT INVENTION
The present invention provides a system that measures distance and velocity
data in real time using time-of-flight (TOF) data rather than relying upon
luminosity data. The system is CMOS-compatible and provides such
three-dimensional imaging without requiring moving parts. The system may
be fabricated on a single IC containing both a two-dimensional array of
CMOS-compatible pixel detectors that sense photon light energy, and
associated processing circuitry.
In applicant's U.S. Pat. No. 6,323,942 B1 (2001) entitled "CMOS-Compatible
Three-Dimensional Image Sensor IC", a microprocessor on a CMOS-compatible
IC continuously triggered a preferably LED or laser light source whose
light output pulses were at least partially reflected by points on the
surface of the object to be imaged For good image resolution, e.g., a cm
or so, a large but brief pulse of optical energy was required, for
example, a peak pulse energy of perhaps 10 W, a pulse width of about 15
ns, and a repetition rate of about 3 KHZ. While average energy in
applicant's earlier system was only about 1 mW, the desired 10 W peak
power essentially dictated the use of relatively expensive laser diodes as
a preferred energy light source. Each pixel detector in the detector array
had associated electronics to measure time-of-flight from transmission of
an optical energy pulse to detection of a return signal. In that
invention, the transmission of high peak power narrow energy pulses
required the use of high bandwidth pixel detector amplifiers.
Applicants' referenced co-pending parent application disclosed a system
that transmitted high frequency component periodic signals having low
average power and low peak power, e.g., tens of mW rather than watts. For
ease of analysis, optical energy periodic signals such as an ideal
sinusoid waveform, e.g., cos(.omega..multidot.t), were assumed, and will
be assumed herein. Emitting such low peak power high frequency component
periodic signals permitted use of inexpensive light sources and simpler,
narrower bandwidth pixel detectors. Bandwidths could be on the order of a
few hundred KHz with an operating (emitted energy modulation) frequency of
about 200 MHz. Good resolution accuracy was still obtainable using a low
peak power optical emitter in that the effective duty cycle is greater
than the output from a narrow-pulsed optical emitter of higher peak power.
In such system and in the present invention, assume that the energy emitted
from the optical source is approximately S.sub.1
=K.multidot.cos(.omega..multidot.t) where K is an amplitude coefficient,
.omega.=2.PI.f, and frequency f is perhaps 200 MHz. Assume further that
distance z separates the optical energy emitter from the target object.
For ease of mathematical representation, K=1 will be assumed although
coefficients less than or greater than one may be used. The term
"approximately" is used in recognition that perfect sinusoid waveforms can
be difficult to generate. Due to the time-of-flight required for the
energy to traverse distance z, there will be a phase shift .phi. between
the transmitted energy and the energy detected by a photo detector in the
array, S.sub.2 =A.multidot.cos(.omega..multidot.t+.phi.). Coefficient A
represents brightness of the detected reflected signal and may be measured
separately using the same return signal that is received by the pixel
detector.
The phase shift .phi. due to time-of-flight is:
.phi.=2.multidot..omega..multidot.z/
C=2.multidot.(2.multidot..PI..multidot.f).multidot.z/C
where C is the speed of light 300,000 Km/sec. Thus, distance z from energy
emitter (and from detector array) is given by:
z=.phi..multidot.C/2.multidot..omega.=.phi..multidot.C/
{2.multidot.(2.multidot..PI..multidot.f)}
Distance z is known modulo 2.PI.C/(2.multidot..omega.)=C/(2.multidot.f). If
desired, several different modulation frequencies of optically emitted
energy may be used, e.g., f.sub.1, f.sub.2, f.sub.3 . . . , to determine z
modulo C/(2.multidot.f.sub.1), C/(2.multidot.f.sub.2),
C/(2.multidot.f.sub.3). The use of multiple different modulation
frequencies advantageously can reduce aliasing. If f.sub.1, f.sub.2,
f.sub.3 are integers, aliasing is reduced to the least common multiplier
of f.sub.1, f.sub.2, f.sub.3, denoted LCM(f.sub.1, f.sub.2, f.sub.3). If
f.sub.1, f.sub.2, f.sub.3 are not integers, they preferably are modeled as
fractions expressible as a.sub.1 /D, a.sub.2 /D, and a.sub.3 /D, where i
in a.sub.i is an integer, and D=(GCD) represents the greatest common
divisor of a.sub.1, a.sub.2, a.sub.3. From the above, distance z may be
determined modulo LCM(a.sub.1, a.sub.2, a.sub.3)/D. This same analytical
approach is also practiced with the various embodiments of the present
invention, described later herein.
Phase .phi. and distance z were determined by mixing (or homodyning) the
signal detected by each pixel detector S.sub.2
=A.multidot.cos(.omega..multidot.t+.phi.) with the signal driving the
optical energy emitter S.sub.1 =cos(.omega..multidot.t). The mixing
product S.sub.1.multidot.S.sub.2 is
0.
5.multidot.A.multidot.{cos(2.multidot..omega..multidot.t+.phi.)+cos(.phi.)
} and will have a time average value of 0.5.multidot.A.multidot.cos(.phi.).
If desired, the amplitude or brightness A of the detected return signal
may be measured separately from each pixel detector output.
To implement homodyne determination of phase .phi. and distance z, each
pixel detector in the detector array had its own dedicated electronics
that included a low noise amplifier to amplify the signal detected by the
associated pixel detector, a variable phase delay unit, a mixer, a lowpass
filter, and an integrator. The mixer mixed the output of low noise
amplifier with a variable phase delay version of the transmitted
sinusoidal signal. The mixer output was lowpass filtered, integrated and
fedback to control phase shift of the variable phase delay unit. In the
equilibrium state, the output of each integrator will be the phase
.psi.(where .psi.=.phi..+-..PI./2) associated with the TOF or distance z
between the associated pixel detector and a point a distance z away on the
target object. The analog phase information is readily digitized, and an
on-chip microprocessor can then calculate z-values from each pixel
detector to an associated point on the target object. The microprocessor
further can calculate dz/dt (and/or dx/dt, dy/dt) and other information if
desired.
However applicants' referenced parent co-pending provisional application
substantially enhances detection sensitivity for such systems in which low
peak power high frequency component periodic signals were used, and in
which phase delay is used to determine TOF, dz/dt (and/or dx/dt, dy/dt,
and other information. More specifically, an improved mixer is described,
in which mixing results from modulating quantum efficiency (QE) of the
photodiodes in the detector array, for example through use of a MOS
transistor gate or altering reverse bias of the photodiodes. Such mixing
offers many advantages including improved high frequency sensitivity,
improved detection signal/noise, smaller form factor, lower power
consumption, and less cost to fabricate.
Several embodiments of QE modulation are described in the present
invention. Conceptually the embodiments may be grouped into two general
categories. One category involves variable phase delay approaches (not
unlike those described in applicants' co-pending application Ser. No.
09/876,373) but in which dedicated electronic mixers (e.g., Gilbert cells)
are replaced by QE modulation. A second category involves mixing with
fixed phase delays using QE modulation, and implements a variety of
spatial and temporal multiplexing approaches. Advantageously, both methods
can modulate QE of MOS-implemented photodiodes by changing photodiode
reverse bias, or by providing MOS-implemented photodiodes with a
photogate, and then changing the gate voltage. Single-ended or
double-ended differential signal processing may be employed with both
methods. Differential QE modulation advantageously allows faster QE
modulation, and provides a differential output that substantially removes
common mode effects due to ambient light and photodiode dark current. In
general, both categories of methods advantageously accumulate
photodetector signal charge on a photodiode capacitor. If desired,
accumulated charge may be examined periodically when QE modulation is
stopped. Such signal accumulation approaches are preferred over methods
that seek to directly measure a high frequency small magnitude
photocurrent.
Using variable phase delay (category one), photocurrent from each
QE-modulated pixel photodiode (or photogate photodiode) is coupled as
input to an associated relatively high input impedance amplifier that need
not exhibit high bandwidth, high frequency response, or high closed-loop
gain. The amplifier output feeds directly to a low pass filter (LPF) whose
output drives an integrator. The integrator output is coupled as to
control phase of the variable phase delay (VPD) that controls QE
modulation signals that drive the photodetector diodes. The VPD is also
driven by a signal from the periodic signal generator that controls the
optical energy emitter. There may or may not be a DC offset associated
with the output signal from the pixel photodiode detectors and with the
homodyne drive signal. Assuming no offsets, at steady-state the LPF output
will be zero. Assuming appropriate DC offsets, at steady-state the LPF
output will be a minima or a maxima. This method may be implemented
single-ended, or preferably double-ended using a complementary approach in
which positive and negative signals are derived from photodiodes that are
QE modulated out of phase.
Using fixed phase delay (category two) fixed homodyne signals are used to
QE modulate each photodetector. In category two, different groups or banks
of photodiode detectors may be defined in a non-localized manner within
the array. For example, a first bank of photodiode detectors may be QE
modulated with fixed 0.degree. phase shift, a second bank may be QE
modulated with fixed 90.degree. phase, shift, a third bank with fixed
180.degree. phase shift, and a fourth bank with fixed 270.degree. phase
shift. Within each pixel, there may be photodiode detectors that
correspond to every one of the four banks. Phase information and target
object brightness information can be determined by examining output values
for each bank within a pixel. This fixed delay approach simplifies the
electronic circuitry associated with each pixel, reduces power
consumption, can reduce IC chip area requirement, and enables a range of
techniques for temporal and spatial multiplexing.
In the various embodiments of the present invention, on-chip measurement
information may be output in random rather than sequential order, and
on-chip signal processing for object tracking and other information
requiring a three-dimensional image can be readily accomplished. The
overall system is small, robust, requires relatively few off-chip discrete
components, and exhibits improved detection signal characteristics.
On-chip circuitry can use such TOF data to readily simultaneously measure
distance and velocity of all points on an object or all objects in a
scene.
Other features and advantages of the invention will appear from the
following description in which the preferred embodiments have been set
forth in detail, in conjunction with their accompanying drawings.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a diagram showing a generic luminosity-based range finding
system, according to the prior art;
FIG. 2A depicts a transmitted periodic signal with high frequency
components transmitted by the present invention, here an ideal cosine
waveform;
FIG. 2B depicts the return waveform with phase-delay for the transmitted
signal of FIG. 2A, as used by the present invention;
FIG. 2C depicts a return waveform similar to that shown in FIG. 2B, but
with a DC-offset level, as used by the present invention;
FIG. 2D depicts a pulse-type periodic waveform of emitted optical energy,
such as might be emitted by a system according to applicants' earlier
invention, now U.S. Pat. No. 6,323,942 B1;
FIG. 2E depicts a non-pulse periodic waveform of emitted optical energy,
according to the present invention;
FIG. 3 is a block diagram of a preferred implementation of the present
invention;
FIG. 4 is a block diagram showing two pixel detectors with their associated
electronics, according to applicants' parent utility application;
FIGS. 5A and 5B are cross-sectioned perspective views of a photodetector
diode, showing reverse bias voltage modulation of depletion layer width to
implement QE modulation, according to the present invention;
FIGS. 6A and 6B depict a photogate photodiode that may be QE modulated by
varying gate voltage, according to the present invention;
FIG. 6C depicts approximate equivalency between an MOS-type photodiode
series-coupled to a capacitor, and a photogate photodiode such as shown in
FIG. 6A, according to the present invention;
FIGS. 7A and 7B depict the equivalent circuit and voltage bias
configurations for the exemplary photodiode of FIGS. 5A and 5B and show,
respectively, high-side and low-side QE modulation, according to the
present invention;
FIG. 7C is a cross-section of an exemplary photodetector structure
illustrating how photon-energy created charges may be recovered using
current, according to the present invention;
FIG. 7D is a cross-section of an exemplary photodetector structure showing
smooth or discrete variation of epitaxial layer dopant concentration,
illustrating how photon-energy created charges may be recovered using
current, according to the present invention;
FIGS. 8A and 8B are side cross-sectional views of two adjacent photodiodes
with a leakage-reducing gate QE modulated 180.degree. out of phase,
according to the present invention;
FIG. 8C is a top view of an array of photodiodes wherein modulation nodes
for alternating banks of photodiodes are coupled in parallel for QE
modulated complementarily to the remaining banks of photodiodes, according
to the present invention;
FIG. 9A is a block diagram showing two photodetectors and their associated
electronics in a single-ended variable phase delay (VPD) QE modulated
embodiment of the present invention;
FIG. 9B is a block diagram of a VPD embodiment showing two pixels
comprising four photodetectors with their associated electronics in which
photodiodes are QE differentially modulated, according to the present
invention;
FIG. 9C is a block diagram of a VPD embodiment showing two pixels
comprising four photodetectors with their associated simplified
electronics including digital integrators, in which photodiodes are QE
differentially modulated, according to the present invention;
FIG. 10 is a block diagram showing two pixels comprising four
photodetectors with their associated electronics in which selectable fixed
phase QE modulation of the photodiodes is used, according to the present
invention;
FIGS. 11A and 11B depict use of tuned inductors with photodiodes in the
configuration of FIG. 10, to reduce power consumption, according to the
present invention;
FIG. 12A is a plan view of a 0.degree.-90.degree.-180.degree.-270.degree.
spatial-multiplexing QE modulation embodiment, showing four adjacent
photodetectors according to the present invention;
FIG. 12B depicts sharing of photodetectors across different pixels for the
spatial-multiplexing QE modulation embodiment of FIG. 12A, according to
the present invention;
FIG. 12C depicts a 0.degree.-120.degree.-240.degree. spatial-division
multiplexing QE modulation embodiment showing three photodetectors,
according to the present invention;
FIGS. 13A and 13B depict differential and single-ended signal processing of
photodetector output, according to the present invention; and
FIGS. 14A and 14B depict circuit configurations to reduce effects of
non-uniform illumination and 1/f noise effects upon photodetectors,
according to the present invention.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS
The present invention advantageously transmits and detects optical energy
that is periodic with a high frequency component, and relies upon phase
shift between transmitted and detected waveforms to discern time-of-flight
and thus z-distance data. Although pulsed-type periodic waveforms may be
used, the present invention will be described with respect to the emission
and detection of sinusoidal waveforms, as such waveforms are rather easily
analyzed mathematically. However it is to be understood that periodic
pulsed waveforms with a high frequency component including imperfect
sinusoidal waveforms are representable mathematically as groupings of
perfect sinusoidal waveforms of varying coefficients and frequency
multiples. The transmission and detection of such waveforms can
advantageously permit use of relatively inexpensive low peak-power optical
emitters, and the use of relatively lower bandwidth amplifiers. This is in
contrast to applicant's referenced U.S. Pat. No. 6,323,942 B1 (2001) in
which a low duty cycle pulse train of narrow pulse widths was emitted by a
very high peak power optical emitter.
FIG. 2A depicts the high frequency component of an exemplary idealized
periodic optical energy signal as emitted by the present invention, here a
signal represented as cos(.omega.t). The period T of the waveform shown is
T=2.multidot..PI./.omega.. The signal is depicted as though it were
AC-coupled in that any magnitude offset is not present. As described
below, the operative frequency of the transmitted signal preferably is in
the few hundred MHz range, and the average and the peak transmitted power
may be relatively modest, e.g., less than about 50 mW or so.
A portion of the transmitted energy reaches a target object and is at least
partially reflected back toward the present invention, to be detected.
FIG. 2B depicts the returned version of the transmitted waveform, denoted
A.multidot.cos(.omega.t+.phi.), where A is an attenuation coefficient, and
.phi. is a phase shift resulting from the time-of-flight (TOF) of the
energy in traversing the distance from the present invention to the target
object. Knowledge of TOF is tantamount to knowledge of distance z from a
point on the object target, e.g., target 20, to the recipient pixel
detector in the array of detectors within a system according to the
present invention.
FIG. 2C is similar to what is shown in FIG. 2B except that in the present
invention, a DC-offset is present. The waveform shown in FIG. 2B may be
described as 1+A.multidot.cos(.omega.t+.phi.). As described later herein,
a DC-offset is desirable in some embodiments for biasing the photodiodes,
but does not really affect the underlying mathematics. Again it is
understood that the period T of the waveform in FIG. 2C, as in FIGS. 2A
and 2B is T=2.multidot..PI./.omega..
FIGS. 2D and 2E are useful in understanding the notion of duty cycle, as
used herein. In a pulse-type periodic signal such as shown in FIG. 2D,
duty cycle d may be defined as the ratio of time T.sub.H /T, where T.sub.H
is the time the signal is higher than a given threshold V.sub.H, and T is
the signal period. Threshold level V.sub.H is usually the average of the
maximum and the minimum signal levels. Within the context of the present
invention, the above definition is analogous, except that T.sub.H will
represent the time during which a photodiode detector 240-x is modulated,
where T is the repetition period associated with turning modulation on and
off for emitter 220, as shown in FIG. 2E. Within the context of the
present invention, the ratio T.sub.H /T can be decreased, provided that
the peak power emission of optical energy emitter 220 is adjusted
appropriately, so as to keep the average power constant. As noted, while
the optical energy emitted by emitter 220 will be periodic, it need not be
a square-wave or square-wave like. A waveform such as shown in FIG. 2E
could be emitted and detected. However it is understood that the above
definitions of duty cycle are also applicable to waveforms such as shown
in FIG. 2E.
Specifying a repetition rate of the transmitted periodic optical energy
signal involves tradeoffs that include considerations of the transmitted
waveshape and duty cycle, the desired granularity in resolving z-distance,
and peak power requirements for the optical energy emitter. For example, a
transmitted periodic signal whose high frequency component is a few
hundred MHz, e.g., 200 MHz, is consistent with z-distance resolution on
the order of a cm or so, assuming eight-bit analog-to-digital conversion
of the detected phase shift information. In practice, assuming a
continuous sinusoidal-type waveform, the peak power required from the
optical energy emitter will be about 10 mW. Of course if the transmitted
waveform duty cycle were decreased to say 1%, the optical energy emitter
peak power would have to be increased to about 500 mW, and so on. It will
be appreciated that the ability to use a low peak power optical emitter is
one of the distinguishing factors between the present invention and
applicant's above-referenced U.S. Pat. No. 6,323,942 B1 (2001).
The processing and use of phase shift information in the present invention
will now be described with reference to FIG. 3, a block diagram depicting
the present invention 200, a three-dimensional imaging system that
preferably is fabricated on a single IC 210. System 200 requires no moving
parts and relatively few off-chip components. Although FIG. 3 is taken
from applicants' referenced co-pending utility patent application, it can
be used to describe the present invention, although circuit details of
various elements in FIG. 3 will be different. In overview, in the various
embodiments of the present invention, preferably each photodetector 240-x
within array 230 has associated electronics 250-x that implements QE
modulation in the photodetectors. Whether using variable phase delay or
fixed phase delay techniques, the present invention determines distance z
according to
z=.phi..multidot.C/2.multidot..omega.=.phi..multidot.C/
{2.multidot.(2.multidot..PI..multidot.f)}.
System 200 includes an optical emitter, for example a low peak power laser
diode, or low peak power LED, that can output a periodic signal with 50 mW
or so peak power when driven with a repetition rate of a few hundred MHz
and, in the preferred embodiment, a duty cycle close to 100%, as duty
cycle is defined herein. At present, useful optical emitters are made from
materials such as AlGaAs, whose bandgap energies are quite different than
that of silicon, from which CMOS IC 210 is preferably fabricated. Thus,
while FIG. 3 depicts optical emitter 220 as being off-chip 210, the
phantom lines surrounding emitter 220 denote that an optical emitter 220
made of CMOS-compatible materials may instead be fabricated on IC 210.
Light source 220 is preferably a low peak power LED or a laser that emits
energy with a wavelength of perhaps 800 nm, although other wavelengths
could instead be used. Below 800 nm wavelength, emitted light starts to
become visible and laser fabrication becomes more difficult. Above 900 nm
CMOS/silicon photodiode efficiency drops off rapidly, and in any event,
1100 nm is the upper wavelength for a device fabricated on a silicon
substrate, such as IC 210. By using emitted light having a specific
wavelength, and by filtering out incoming light of different wavelength,
system 200 can operate with or without ambient light. The ability of
system 200 to function in the dark can be advantageous in certain security
and military type imaging applications. Off-chip mounted lens 290
preferably focuses filtered incoming light energy onto sensor array 230
such that each pixel detector 240x receives light from only one particular
point (e.g., an object surface point) in the field of view. The properties
of light wave propagation allow an ordinary lens 290 to be used to focus
the light onto the sensor array. If a lens (290') is required to focus the
optical light energy transmitted from emitter 220, a single lens could be
used for 290, 290' if a mirror-type arrangement were used. Typical LED or
laser diode emitters 220 have a shunt capacitance of perhaps 100 pF. Thus
in driving emitter 220, it would be advantageous to place a small
inductance (perhaps a few nH) in parallel with this capacitance, where the
combined inductance-capacitance resonate at the periodic frequency of the
emitter, typically a few hundred MHz. Alternatively, inductance (again a
few nH) can be series-coupled to the emitter and its parasitic
capacitance. If desired, such inductance can be derived using a bonding
wire to the emitter.
CMOS-compatible IC 210 will preferably have fabricated thereon oscillator
225 driver, array 230 (comprising perhaps 100.times.100 (or more) pixel
detectors 240 and 100.times.100 (or more) associated electronic processing
circuits 250), microprocessor or microcontroller unit 260, memory 270
(which preferably includes random access memory or RAM and read-only
memory or ROM), and various computing and input/output (I/O) circuitry
280, including, for example an analog/digital (A/D) conversion unit
providing perhaps 8-bit A/D conversions of phase information .phi.
detected by the various pixel detectors in array 230. Depending upon
implementation, a single on-chip A/D converter function could be provided,
or a dedicated A/D converter could be provided as part of each electronic
processing circuit 250. I/O circuit 280 preferably can also provide a
signal to control frequency of the oscillator 225 that drives the energy
emitter 220.
The DATA output line shown in FIG. 3 represents any or all information that
is calculated by the present invention using phase-shift information from
the various pixel detectors 240 in array 230. Preferably microprocessor
260 can examine consecutive frames stored in RAM 270 to identify objects
in the field of view scene. Microprocessor 260 can then compute z-distance
and can compute object velocity dz/dt, dx/dt, dy/dt. Further,
microprocessor 260 and associated on-chip circuitry can be programmed to
recognize desired image shapes, for example a user's fingers if an
application using system 200 to detect user interface with a virtual input
device. The data provided by microprocessor 260 could be reduced to
keystroke information in such an application. Any or all of this data
(denoted DATA in FIG. 3) can be exported from the IC to an external
computer for further processing, for example via a universal serial bus.
If microprocessor 260 has sufficient computational power, additional
on-chip processing may occur as well. Note too that output from the array
of CMOS-compatible detectors 240 may be accessed in a random manner if
desired, which permits outputting TOF DATA in any order.
Among its other functions, microprocessor 260 acting through interface
circuit 280 causes driver 225 to oscillate periodically with a desired
duty cycle at a desired frequency, for example f.sub.1 =200 MHz. In
response to signals from oscillator driver 225, laser diode or LED 220
emits optical energy at the desired frequency, e.g., f.sub.1 =200 MHz and
duty cycle. Again, while a sinusoid or cosine waveform is assumed for ease
of mathematical representation, a periodic waveform with similar duty
cycle, repetition rate and peak power may be used, e.g., perhaps
squarewaves. As noted, average and peak power is advantageously quite
modest in the present invention, for example 10 mW. As a result, the cost
of an LED optical emitter 220 is perhaps 30.cent. compared to a cost of
many dollars for a high peak power laser diode in applicant's earlier
invention, described in U.S. Pat. No. 6,323,942 B1 (2001).
The optical energy whose periodic high frequency component is ideally
represented as S.sub.1 =cos(.omega.t) is focused by optional lens 290'
upon target object 20, some distance z away. At least some of the optical
energy falling upon target 20 will be reflected back towards system 200
and will be detected by one or more pixel detectors 240 in array 230. Due
to the distance z separating system 200, more particularly a given pixel
detector 240 in array 230, and the target point on object 20, the detected
optical energy will be delayed in phase by some amount .phi. that is
proportional to time-of-flight, or to the separation distance z. The
incoming optical energy detected by different pixel detectors 240 can have
different phase .phi. since different times-of-flight or distances z are
involved. In various figures including FIG. 3, the incoming optical energy
is denoted as S.sub.2 =A.multidot.cos(.omega.t+.phi.), e.g., the AC
component of a return signal that will in fact include a DC component.
However the DC component is relatively unimportant and is not depicted in
the figures.
As will be described, it is the function of electronics 250 associated with
each pixel detector 240 in array 230 to examine and determine the relative
phase delay, in cooperation with microprocessor 260 and software stored in
memory 270 executed by the microprocessor. In an application where system
200 images a data input mechanism, perhaps a virtual keyboard,
microprocessor 260 may process detection data sufficient to identify which
of several virtual keys or regions on a virtual device, e.g., a virtual
keyboard, have been touched by a user's finger or stylus. Thus, the DATA
output from system 200 can include a variety of information, including
without limitation distance z, velocity dz/dt (and/or dx/dt, dy/dt) of
object 20, and object identification, e.g., identification of a virtual
key contacted by a user's hand or stylus.
Preferably IC 210 also includes a microprocessor or microcontroller unit
260, memory 270 (which preferably includes random access memory or RAM and
read-only memory or ROM), and various computing and input/output (I/O)
circuitry 280. For example, an output from I/O circuit 280 can control
frequency of the oscillator 225 that drives the energy emitter 220. Among
other functions, controller unit 260 may perform z distance to object and
object velocity (dz/dt, dy/dt, dx/dt) calculations. The DATA output line
shown in FIG. 3 represents any or all such information that is calculated
by the present invention using phase-shift information from the various
pixel detectors 240. The two-dimensional array 230 of pixel sensing
detectors preferably is fabricated using standard commercial silicon
technology. This advantageously permits fabricating a single IC 210 that
includes the various pixel detectors 240 and their associated circuits
250, as well as circuits 225, 260, 270, 280, and preferably the energy
emitter 220 as well. Understandably, the ability to fabricate such
circuits and components on the same IC with the array of pixel detectors
can shorten processing and delay times, due to shorter signal paths. In
FIG. 3, while system 200 may include focusing lens 290 and/or 290', it is
understood that these lenses will be fabricated off IC chip 210.
Each pixel detector 240 is equivalent to a parallel combination of a
current source, an ideal diode, shunt impedance, and noise current source,
and will output a current proportional to the amount of incoming photon
light energy falling upon it. Preferably CMOS fabrication is used to
implement the array of CMOS pixel diodes or photogate detector devices.
Exemplary photodiode fabrication techniques include diffusion-to-well,
diffusion-to-substrate, a well-to-substrate junction, and photogate
structures. Well-to-substrate photodiodes are more sensitive to infrared
(IR) light, exhibit less capacitance, and are thus preferred over
diffusion-to-substrate photodiodes.
As noted FIG. 4 represents an embodiment described in applicants'
co-pending utility patent application. FIG. 4 represents a portion of IC
210 and of array 230, and depicts pixel detectors 240-1 through 240-x, and
each diode's associated exemplary electronics 250'-1 through 250'-x. For
ease of illustration in various figures including FIG. 4, lens 290 is not
depicted. FIG. 4 does not relate directly to the present invention, but is
included to provide a better understanding and appreciation for the
benefits provided by the present invention. In the description that
follows, FIGS. 9A-9C are directed to category one VPD QE modulation
techniques and FIGS. 10A-10C are directed to category two fixed phase
modulation techniques, with the remaining figures illustrating aspects of
these various techniques.
In FIG. 4, only two pixel diodes 240 and two associated electronic circuits
250' are depicted, for ease of illustration however an actual array will
include hundreds or thousands or more of such pixel detectors and
associated electronic circuits. As noted, if desired a dedicated A/D
converter could be provided as part of each electronics circuit 250'-1
through 250'-x, as opposed to implementing an omnibus A/D function on IC
chip 210.
Consider now detection of incoming optical energy by pixel detector 240-1.
Assuming that a low power LED or laser diode or the like 220 emits optical
radiation having idealized high frequency component S.sub.1
=cos(.omega..multidot.t), a fraction of such radiation reflected from a
point on the surface of target 20 (distance z away) is given by S.sub.2
=A.multidot.cos(.omega..multidot.t+.phi.). Upon receiving this incoming
radiation, pixel detector 240-1 outputs a signal that is amplified by low
noise amplifier 300. An exemplary amplifier 300 might have a closed-loop
gain of perhaps 12 dB.
As noted, periodic emissions from optical source 220 preferably are
sinusoidal or sinusoidal-like with a high frequency component of a few
hundred MHz. Despite this high optical modulation frequency, it suffices
for amplifier 300 to have a bandwidth of perhaps 100 KHz or so, perhaps as
low as tens of KHz because all frequencies of interest are close to this
modulation frequency. It will be appreciated that providing hundreds or
thousands of low noise, relatively low bandwidth amplifiers 300 on IC 210
is an easier and more economical undertaking than providing high bandwidth
amplifiers able to pass narrow pulses, as in applicant's parent invention.
Thus, in FIG. 4, array 230 can function with relatively small bandwidth
amplifiers 300, where each amplifier output is coupled directly to a first
input of an associated mixer 310, whose second input is a signal of like
frequency as that present at the first input. If each amplifier 300 and
its associated mixer 310 were implemented as a single unit, it could
suffice for the overall unit to have a bandwidth on the order of tens of
KHz, and a high frequency response also on the order of tens of KHz.
As shown in FIG. 4, when comparing the detected signal to the transmitted
signal, there will be a phase shift .phi. that is related to TOF and to
distance z. Each circuit 250'-x couples the output of the associated low
noise amplifier 300 to the first input of a mixer 310. In applicants'
earlier invention for which FIG. 4 is descriptive, mixer 310 could be
implemented as Gilbert cells, multipliers, etc.
In essence, each mixer 310 will homodyne the amplified detected output
signal S.sub.2 from an associated pixel detector 240 with a generator 225
signal S.sub.1. Assuming that the optical energy emitted has an idealized
high frequency component represented as a sine wave or cosine wave, the
mixer output product S.sub.1.multidot.S.sub.2 will be
0.
5.multidot.A.multidot.{cos(2.multidot..omega..multidot.t+.phi.)+cos(.phi.)
} and will have an average value of 0.5.multidot.A.multidot.cos(.phi.). If
desired, the amplitude or brightness A of the detected return signal may
be measured separately from each pixel detector output. In practice, an
eight-bit analog-to-digital resolution of A.multidot.cos(.phi.) will
result in about 1 cm resolution for z-measurements.
Each mixer 310 will have a second input coupled to the output of a variable
phase delay (VPD) unit 320. VPD units 320 may be implemented in many ways,
for example using a series-coupled string of inverters whose operating
power supply voltage is varied to speed-up or slow-down the ability of
each inverter to pass a signal. A first input to each VPD unit 320 will be
derived from signal generator 225, and will be S.sub.1 =cos(.omega.t),
give or take a signal coefficient. Assume that VPD 320 adds a variable
time delay .psi. to the cos(.omega.t) signal derived from generator 225.
Mixer 310 then mixes the amplified cos(.omega..multidot.t+.phi.) signal
output by amplifier 300 with the cos(.omega..multidot.t+.psi.) signal
output by VPD 320. Mixer 310 now outputs signals including
0.
5.multidot.A.multidot.{cos(.phi.-.psi.)+cos(2.multidot..omega..multidot.t+
.phi.+.psi.)}. The output of mixer 310 is coupled to the input of a low
pass filter 340 that preferably has a bandwidth of a 100 Hz or so to a few
KHz or so, such that the output from filter 340 will be a low frequency
signal proportional to 0.5.multidot.A.multidot.cos(.phi.-.psi.). This low
frequency signal is now input to an integrator 330 whose output will be
.phi..sub.x for pixel detector 240-x.
VPD 320 is driven by two signals that each have the same modulation
frequency as that emitted by optical emitter 220, albeit with a phase
difference (.phi.-.psi.). Note that if phase shift
.psi.=.phi..+-.90.degree., the polarity of integrator 330 output will
change. In the configuration shown in FIG. 4, phase shift .psi..sub.x
=.phi..sub.x.+-.90.degree. associated with the return signal detected by
each pixel detector 240-x is available from that pixel detector's
integrator 330-x.
The phase shift .phi. due to time-of-flight may be given by:
.phi.=2.multidot..omega..multidot.z/
C=2.multidot.(2.multidot..PI..multidot.f).multidot.z/C
where C is speed of light 300,000 Km/sec. Thus, distance z from energy
emitter 220 to a pixel detector 240-x in array 230 is given by:
z=.phi..multidot.C/2.multidot..omega.=.phi..multidot.C/
{2.multidot.(2.multidot..PI..multidot.f)}
Distance z is known modulo 2.PI.C/(2.multidot..omega.)=C/(2.multidot.f).
Using several different modulation frequencies such as f.sub.1, f.sub.2,
f.sub.3 . . . , permits determining distance z modulo
C/(2.multidot.f.sub.1), C/(2.multidot.f.sub.2), C/(2.multidot.f.sub.3),
etc., and further avoids, or at least reduces, aliasing. For example,
microprocessor 260 can command generator 225 to output sinusoidal drive
signals of chosen frequencies, e.g., f.sub.1, f.sub.2, f.sub.3, etc. If
f.sub.1, f.sub.2, f.sub.3 are integers, e.g., i=integer, aliasing is
reduced to the least common multiplier of f.sub.1, f.sub.2, f.sub.3,
denoted LCM(f.sub.1, f.sub.2, f.sub.3). If f.sub.1, f.sub.2, f.sub.3 are
not integers, they preferably are modeled as fractions expressible as
a.sub.1 /D, a.sub.2 /D, a.sub.3 /D, where a.sub.i denotes integer i, and
D=GCD(a.sub.1, a.sub.2, a.sub.3), where GCD denotes greatest common
divisor. Distance z can then be determined modulo LCM(a.sub.1, a.sub.2,
a.sub.3)/D.
The closed-loop feedback circuit configuration of FIG. 4 reaches a stable
point when the two input signals to each mixer 310 are 90.degree. out of
phase with respect to each other, e.g., at a chosen one of .psi..sub.x
=.phi..sub.x +90.degree. or .psi..sub.x =.phi..sub.x -90.degree.,
depending upon circuit implementation. At the proper 90.degree.
out-of-phase steady-state, the output signal from each lowpass filter 340
will be, ideally, null. For example, should the output signal from a
lowpass filter 340 signal go positive, then the output signal from the
associated integrator 330 will add more phase shift to drive the lowpass
filter output back towards a null state.
When the feedback system is at a stable state, the pixel detector
electronics 250'-x in array 230 provide various phase angles .psi..sub.1,
.psi..sub.2, .psi.3, . . . .psi..sub.N, where .psi..sub.x
=.phi..sub.x.+-.90.degree.. The phase angles are preferably converted from
analog format to digital format, for example using an analog/digital
converter function associated with electronics 280. If desired,
electronics 250'-x could mix signals having a constant phase value for all
pixels. Advantageously microprocessor 260 can then execute software, e.g.,
stored or storable in memory 270 to calculate z-distances (and/or other
information) using the above mathematical relationships. If desired,
microprocessor 260 can also command generator 225 to output discrete
frequencies e.g., f.sub.1, f.sub.2, f.sub.3. . . to improve system
performance by reducing or even eliminating aliasing errors.
Referring still to FIG. 4, various implementations may be used to generate
phase angle .psi.=.phi..+-.90.degree.. Assume that a given application
requires acquisition of an image at a frame rate of 30 frames/second. In
such application, it suffices to sample phase angle .psi. during A/D
conversion with a sample rate of about 30 ms. This sample rate is
commensurate with the relatively low bandwidth otherwise present within
electronics 250'-x, as shown in FIG. 4. In practice, system 200 can
provide z-distance resolution of about 1 cm and in practical applications,
z-range will be within perhaps 100 m or less.
Although z-distance is determined from TOF information acquired from phase
delay .psi., it is noted that the relative brightness of the signals
returned from target object 20 can also provide useful information. The
amplitude coefficient "A" on the return signal is a measure of relative
brightness. While the feedback configuration of FIG. 4 seeks to achieve a
minimum output signal from the lowpass filters 340, with slight alteration
a maximum lowpass filter output signal could instead be used, the output
signal then representing brightness coefficient A. Such a configuration
could be implemented using a signal 90.degree. out-of-phase with the
output from VPD 320 to modulate another copy of the output of the low
noise amplifier 300. The average amplitude of the thus-modulated signal
would be proportional to coefficient A in the incoming detected return
signal.
Having completed describing applicants' former invention, various
embodiments of the present invention will now be described, primarily with
reference to FIGS. 9A-9C (category one), and FIG. 10 (category two). In
the present invention, dedicated electronic mixers (such as were used in
the earlier invention described herein in FIG. 4) are avoided, and instead
quantum efficiency (QE) modulation techniques are used. These QE
modulation techniques advantageously can accumulate detected signal
charge, and are preferred over methods that attempt to directly measure
high frequency, small magnitude detection photocurrent-generated signals.
Before categorizing QE modulation circuit topologies according to the
present invention, it is useful to describe MOS diode behavior and how MOS
diode quantum efficiency can be varied by bias potential and/or photogate
potential. FIGS. 5A and 5B depict a portion of IC 210 and array 230, and
depict a portion of a single photodiode detector 240, shown here
fabricated on a p doped substrate 410. Photodiode 240 is shown with a
depletion layer 420 having depth W, above which are found lightly doped
and more heavily doped n regions 430 and 440. (The terms depletion layer
and depletion region may be used interchangeably herein.) The n+doped
region 440 serves as the photodiode anode, the connection to which is
shown as 450. A p+doped region 460 formed at the upper region of substrate
420 serves as the photodiode cathode, connection to which is shown as 470.
A depletion region 480 having depletion width W exists between--region 430
and p substrate region 410. (It is understood that doping polarities
described herein may be inverted, and that structures may be fabricated on
n substrate material rather than on the described p substrate material.)
The width W of depletion region 480 will vary or modulate with changes in
reverse bias voltage coupled between the photodiode anode 450 and cathode
470. This bias potential is denoted Vr1 in FIG. 5A, and is denoted Vr2 in
FIG. 5B. In FIGS. 5A and 5B, Vr2>Vr1, with the result that the width W
of the depletion region increases.
Photons representing incoming optical energy, e.g., energy reflected from
target object 20 perhaps, will fall upon photodiodes 240-x in array 230,
e.g., see FIG. 3, among other figures. The photons can generate
electron-hole pairs in the depletion region of these photodiodes and also
in the quasi-neutral regions. These electron-hole pairs have a relatively
long lifetime before recombining. Photons that generate electron-hole
pairs in the depletion region advantageously have a much higher per photon
photoelectric current contribution than photons that generate
electron-hole pairs in the quasi-neutral regions of the substrate. This is
because electron-hole pairs generated in the depletion region are quickly
swept away by the electric field, and will strongly contribute to the
resultant photocurrent. By contrast, electron-hole pairs generated in the
quasi-neutral region remain there for some time and experience a greater
probability of recombination without making substantial contribution to
the photocurrent. It is seen that increasing the depletion region width W
provides a larger region in which electron-hole pairs may be created and
quickly swept away to contribute to the photocurrent, thus enhancing the
quantum efficiency of the photodiode.
Those skilled in the relevant art will recognizes that depletion width W
may be expressed as:
W=[2.epsilon..multidot.(.psi..sub.0 =V.sub.R -V.sub.B)].sup.0.5
{[qN.sub.A.multidot.(1+N.sub.A /N.sub.D)].sup.-0.5
+[qN.sub.D.multidot.(1+N.sub.D /N.sub.A)].sup.-0.5 }
where (V.sub.R -V.sub.B) is the reverse bias of photodiode 240, N.sub.A and
N.sub.d are respective doping concentrations for the diode n and p
regions, and .psi..sub.0 =V.sub.T In(N.sub.A N.sub.D /n.sub.i.sup.2),
where V.sub.T =kT/q=26 mV, and n.sub.i =1.5.multidot.10.sup.10 cm.sup.-3.
Quantum efficiency (QE) modulation according to the present invention
recognizes from the above equation that photodiode depletion width W can
be modulated by varying the reverse bias coupled between the anode and
cathode regions of the photodiode. This in turn permits varying the
quantum efficiency (QE) of the photodiode, which can result in improved
detection sensitivity for the overall system. Table 1 depicts exemplary
data for a discrete PIN photodiode exposed to a fixed level of
illumination, and shows measured photodiode current as a function of
reverse bias voltage coupled to the photodiode. Data for a
CMOS-implemented photodiode may of course differ from what is shown in
Table 1.
TABLE 1
Reverse voltage (VDC) Photodiode current (mA)
0.2 0.09
0.5 0.38
1 0.83
2 1.4
3 1.51
4 1.62
5 1.7
6 1.66
7 1.76
8 1.8
10 1.8
Note in Table 1 that for the exemplary PIN photodiode, magnitude of the
photodiode current (e.g., photocurrent) varies by a factor of four as the
reverse bias is changed between 0.5 VDC and 2 VDC.
Modulating the photodiode reverse bias is a mechanism by which QE can be
varied to improve detection sensitivity of photodiodes in an array.
However, an even more efficient implementation of a QE modulation detector
uses a photogate structure. In such embodiment, the photodiodes preferably
are implemented as photogate MOS photodiodes whose QE is modulated by
varying potential coupled to the gate of the photogate structure.
Referring now to FIGS. 6A and 6B, assume that substrate 410 is p-type
material, and that MOS-type source and drain regions, respectively S and
D, are formed with n-doped material, although as noted earlier doping
polarity types could of course be reversed. Assume too that source S and
drain D are connected together, as shown in FIG. 6A. When the voltage S1
(t) coupled to gate G is high, device 240-x will deplete and then invert,
again assuming an n-channel device. In this configuration, gate G and
underlying thin oxide (TOX) are assumed substantially transparent to
incoming photon energy S2(t). This condition may be met if the polysilicon
material used to form gate G is not polycided.
Referring to FIGS. 6A and 6B, gate structure G is substantially transparent
to incoming optical energy shown as S2(t). The structure shown in FIG. 6A
includes both source and drain regions, denoted S and D. By contrast, the
structure of FIG. 6B is formed without the drain structure, to improve
quantum efficiency modulation. In FIG. 6A, since the source and drain
regions are connected together, device 240x can operate without a drain
region, as shown in FIG. 6B. As noted, MOS fabrication processes
preferably are used to implement IC 70, upon which the present invention
may be implemented. With many MOS fabrication processes, the drain region
of device 240x may be omitted as shown in FIG. 6B. Omitting the drain
region effectively increases relative variation in the device collection
efficiency between the low sensitivity operating state and the high
sensitivity operating state. As described below, changing bias of the
optically transparent gate potential changes shape of the depletion layer:
a layer 480 substantially confined about the source region is present when
the gate bias is low, which depletion layer region 480' extends
substantially under the gate region when the gate bias is high.
Photocharges, e.g., EH1, EH2, etc. are generated in the substrate under the
gate region in response to photon energy S2(t). If no channel exists under
the gate region, then most of the photocharges will be lost, and only the
source and drain regions will collect photocharge. But if the region under
the gate is inverted and/or depleted, then generated photocharges can be
captured and swept into the source and drain regions. This effectively
increases efficiency of the photon collecting structure 240-x. The
increase in collection efficiency is roughly proportional to the ratio of
area under gate G and the area of the source and drain regions, S and D.
If photogate devices 240x are properly sized, this ratio can be 10:1 or
greater. The increase in efficiency occurs abruptly, with the efficiency
suddenly increasing when the voltage S1(t) exceeds a threshold level. If
the channel area is undoped and substrate doping is above 10.sup.17, the
threshold will be about 0 V, such that the photogate photodetector 240x is
in low sensitivity mode at a gate voltage of about -0.1 V and in a high
sensitivity mode when the gate voltage is about +0.1 V. It will be
appreciated that a relatively small change in gate voltage can bring about
a substantial change in sensitivity of the device.
FIG. 6C depicts the approximate circuit equivalency between a photogate
photodiode 240X and a more conventional MOS photodiode D1 coupled to a
capacitor C.sub.O. Understandably, voltage levels for MOS photodiodes may
differ from voltage levels for photogate photodiodes. Thus, it will be
appreciated that the term photodiode or photodetector or pixel detector
240x may be understood to include a photogate photodiode such as described
above with respect to FIGS. 6A-6C. Similarly, the various circuits and
analyses for QE modulation described herein with respect to a more
conventional MOS photodiode may also be understood to be practicable with
a photogate photodiode 240x, such as described above. For ease of
illustration, most of the embodiments herein are described with reference
to a MOS-type photodiode detector rather than a photogate detector,
however either type detector may be used.
FIGS. 7A and 7B depict the equivalent circuit of a photodiode detector 240,
which is denoted D1 and includes a parasitic shunt capacitor C.sub.1. FIG.
7A may be referred to as depicting high-side QE modulation in that the
modulation signal is coupled via capacitor C.sub.O. In FIG. 7B, the
modulation signal is coupled via capacitor C1 and the figure may be said
to depict low side QE modulation. In FIG. 7B, capacitor Co is generally
located within an amplifier (not shown) in the electronics associated with
pixel detector D1.
In the right hand portion of FIG. 7A, an excitation source V2 is coupled to
a light emitter L1, e.g., a laser diode or an LED, so as to cause L1
photoemission that is proportional to V2. In the left hand portion of FIG.
7A, photodiode D1 receives such photon energy from L1, and a photocurrent
l1 is induced in response. It is understood that photodiode D1 (e.g.,
photodiodes 240-x in array 230) will be reverse biased, and bias source V1
will thus include a voltage offset. Alternatively, photodiode node N.sub.d
can be pre-charged during initialization, before detection of an incoming
signal. It will be appreciated that V2 in FIGS. 7A and 7B may be analogous
to periodic waveform generator 225, and that L1 may be analogous to
optical energy emitter 220 (see FIG. among other figures).
In FIGS. 7A and 7B, photodiode reverse bias voltage and hence the QE of the
photodiode is modulated by bias source V1. In FIG. 7A, the reverse bias
voltage is given by Vd1=V1.multidot.(C.sub.0)/(C.sub.0 +C.sub.1), where
C.sub.0 is series-coupled between V1 and D1. From Table 1 and FIGS. 5A and
5B, a large magnitude V1 represents a larger reverse bias that can
advantageously increase the width W of the photodiode depletion region.
This in turn increases sensitivity of photodiode D1 (or 240), with the
result that photodiode current l1 increases in response to incoming photon
energy from L1 (or incoming photon energy reflected from a target object
20).
If excitation source V.sub.2 and bias source V.sub.1 operate at the same
frequency (.omega.), the total charge provided by current source I.sub.1
per cycle is maximized when V.sub.1 and V.sub.2 are in phase, e.g., when
magnitude of V.sub.1 (.omega.t) and V.sub.2 (.omega.t) are high
simultaneously. This results because photodiode sensitivity will be
maximum when incoming photon energy is at the highest magnitude, or
brightest intensity. Conversely, if D1 sensitivity is minimal when the
incoming photon energy is maximum, then the amount of charge sourced per
cycle by I.sub.1 is minimized.
The change in amount of charge .DELTA.Q.sub.N on photodiode node N.sub.d
after a given number of cycles will be the amount of charge sourced by
I.sub.1 during those cycles. The change .DELTA.Q.sub.N can be determined
by measuring the difference in voltage .DELTA.V.sub.D on node N.sub.d
before and after capacitors C.sub.0 and C.sub.1 have been discharged by
the photocurrent I1. Normally photocurrent I1 is very small and difficult
to measure directly. However its accumulated effect over a large number of
cycles results in a measurable voltage change .DELTA.V.sub.D.
If the photodiode anode and cathode terminals can each be set to an
arbitrary voltage in FIG. 5B, then the upper lead of C.sub.0 can be at
ground potential, as shown in FIG. 7B. As described later with respect to
several embodiments, typically node N.sub.d is coupled to an amplifier
input that also has a shunt capacitor coupled to the same input node. An
advantage of the configuration of FIG. 7B is that the parasitic shunt
capacitance of the amplifier can be used as C.sub.1 in lieu of an
additional or dedicated shunt capacitor. So doing can reduce parts count
and reduce the area required to implement the present invention on an IC
chip. Furthermore, this configuration produces less noise and less
susceptibility to variations in production technology.
When photon energy falls upon a photodiode, there is a time lag between
arrival of the incoming photon energy and collection of freed electrons.
This time lag increases substantially with optical energy wavelength, and
can be on the order of a few ns for wavelengths of about 850 nm.
Accordingly, optical energy emitter 225 may be selected to emit smaller
wavelengths such that photodetectors 240-x in array 230 have more rapid
response and may be QF modulated at higher frequency .omega..
Understandably, it is desired that photodetectors used in the various
embodiments of the present invention detect not only efficiently, but
rapidly as well. Use of a light emitter 220 to transmit optical energy of
relatively shorter wavelength can promote detector efficiency, but such
emitters are more expensive to fabricate than emitters that provide longer
wavelength energy. For example a relatively inexpensive laser diode may be
used as emitter 220 to transmit energy of perhaps 850 nm wavelength. While
such an emitter is relatively inexpensive, the longer wavelength will
penetrate more deeply into the structure of the pixel detectors, e.g., at
least 7 .mu.m, with resultant loss of quantum efficiency and slow
response.
Referring now to the exemplary CMOS structure of FIG. 7C, quantum
efficiency suffers because much of the incoming photon energy reflected by
the target object 20 will create electron-hole pairs (EHx) deep within the
epitaxial region 410 of the pixel photodetectors 240, and may also create
electron-hole pairs EHx' more deeply in the structure, in region 412.
Unfortunately, many of these deeply-freed electrons will be unable to
reach the surface region of the photodiode detector where they can be
collected and would thus contribute to the photodiode detection signal
current. Further, use of longer wavelength energy also produces an
undesired time delay before signal current is generated. The delay,
typically a few ns, occurs because diffusion effects predominate over
drift effects in collecting such deeply-freed electrons as may contribute
to the detection photodiode current.
If somehow the electrons associated with EHx, EHx', were moved closer to
surface region of the photodiode structure, then drift effects would
predominate over diffusion effects, and the detection current would be
seen sooner. Because doping of epitaxial layer 410 is very low, it is
possible to move electrons created deep within the epitaxial layer using
relatively small currents.
Referring to FIG. 7C, epitaxial layer 410 is typically on the order of 7
.mu.m thick with a dopant concentration of about N.sub.A =10.sup.15
/cm.sup.3, and underlying heavily doped substrate region 412 is on the
order of several hundred .mu.m thick, and has a dopant concentration of
about N.sub.A =10.sup.18 /cm.sup.3. Structures such as shown in FIG. 7C
are readily available from many commercial vendors.
In FIG. 7C, an n-well region 430 and a p++ region 460 are defined in the
epitaxial layer 410. N+ region 440 is formed with the n-well region 430.
As described below, collection leads 445, 447 are provided to facilitate
moving 20 the deeply-free charges around and preferably in an upward
direction for collection by n-well 430. (It is understood that the dopant
polarities described could be reversed, e.g., an n-type substrate might
instead be used, and that dopant levels and structure thicknesses may also
be modified.)
What will now be described is a method by which charges associated with EHx
may be moved upward to enable their eventual collection by n-well 430 due
to diffusion current, once the charges are in sufficiently close proximity
to the n-well. The goal is to urge deeply-freed electrons upward
sufficiently slowly to be collected by lead 445 associated with the
n-well, but not by lead 447 associated with the p++ region. While the
method to be described can successfully collect electrons associated with
electron-hole pairs EHx, the method cannot reach more deeply into the
structure to also collect electrons associated with EHx'. Such movement is
shown by the phantom right-angle line in FIG. 7C. To attempt to also
recover the EHx' electrons would require an unacceptably large current due
to the high dopant level associated with layer 412.
Consider now magnitude of electrical current required to move electrons
according to the present invention. Assume that, when viewed from the top,
the structure shown in FIG. 7C is a square of dimension 1 .mu.m .times.1
.mu.m, whose area is denoted A.sub.s. For a 7 .mu.m region 410 thickness,
the resultant volume is 7.times.10.sup.-12 /cm.sup.3. The requisite charge
that must be removed from such a volume is
10.sup.15.times.10.sup.-8.times.7.times.10.sup.-4.times.1.6.times.10.sup.
-19 As=1.12.times.10.sup.-15 A.sub.s, where 1.6.times.10.sup.-19 is the
charge associated per electron. If the goal is remove this much charge
within, say, 1 ns, then the required current is about 1.12 .mu.A. While
this current is not negligible, it is indeed feasible to provide this
current for each square micron associated with photodetector array 230.
For an array sized 1 mm.times.1 mm, modulated at 200 MHz, total current
would be on the order of a 200 mA to move electrons upward 7 .mu.m. It
will be appreciated that the high dopant level associated with substrate
region 412 precluded attempting to recover electrons from EHx' using this
method.
Thus, one approach to somehow moving deeply-freed electrons from layer 410
upward for collection is to sweep substantially all holes downward by
about 7 .mu.m. Since electron and hole mobility are reasonably close, such
freed electrons will be moved upward at least 7 microns and can come in
sufficiently close proximity to n-well region 430 to be favorably
influenced by the depletion region therein. The depletion region influence
will promote collection of such deep-freed electrons higher in the
structure.
By establishing a preferably pulsed current below n-well region 430, holes
can be made to move downward by about 7 .mu.m, while electrons will be
made to move upward by at least the same distance due to their higher
mobility. As noted, once the electrons come sufficiently close to be
influenced by the electric field setup by the depletion region within the
n-well region, the likelihood of collecting the electrons can be
substantially enhanced.
In one embodiment, ohmic contact 460 is formed on the substrate outside
n-well region 430 and is used to help bring electrons close to the
depletion layer. This approach can work well in that the epitaxial layer
410 has a relatively low dopant concentration, and the magnitude of charge
required to sweep electrons upward by about 7 .mu.m is acceptable. There
is no incentive to encourage upward movement of electrons by more than
about 7 .mu.m as there would be too many holes in the more heavily doped
regions encountered at the upper levels of structure 210. If desired,
rather than form an ohmic contact, an AC-coupled approach using a
capacitor structure could instead be used.
A detector structure employing various types of epitaxial region doping
gradients will now be described. FIG. 7D depicts a structure that may be
similar to that of FIG. 7C, although the depth of structure 240' in FIG.
7D may be deeper than about 7 .mu.m. In FIG. 7D, the epitaxial layer 410'
preferably defines different dopant concentrations that range from a
relatively high concentration (p1) to a lower concentration (p3). The
dopant concentration transition may be a continuum, or may be more
discrete, e.g., by forming separate epitaxial layers, each having an
associated dopant concentration.
Those skilled in the art will recognize that there exists an electric field
that is associated with each doping region boundary. For structure 240' in
which dopant concentration is weaker nearer the upper surface of the
structure, the direction of the electric field may be defined as being
downward. Electrons in EHx' near the upper surface of region 412 will move
upward through the interface existing between regions 412 and p1 due to
the electric field at that interface. Since these electrons will not move
downward through that interface, there is an excellent probability that
they can be induced to quickly move upward (by diffusion effects) close to
the next epitaxial doping interface (p1, p2), from whence they can again
be induced to move into the next dopant region, here p2, due to the
electric field existing at p1, p2. Once in that (less highly doped)
epitaxial region (here, p2) the electrons again will no longer move
downward through the p1, p2 interface, and have an excellent chance of
moving upward to be influenced by the next epitaxial region (p3), from
whence they can be induced to move into that region, and so forth.
Understandably the same above-described phenomenon works for electrons
initially from pairs EHx that were initially freed somewhere in the
epitaxial region. It is also understood that fewer or more than three
dopant concentrations or regions may be defined within the epitaxial
region.
Thus, a drift current phenomenon associated with the electric fields in the
various p1, p2, p3, . . . interface or boundary regions comprising the
epitaxial layer induces the electrons to move quickly upwards through each
of the p1, p2, . . . interface regions.
As above-described, discretely doped epitaxial regions serve somewhat as
"staging" or "holding" regions for electrons that have come sufficiently
close to be moved into the region. However if a continuum of dopant
gradient can be defined throughout the epitaxial region 410', there would
be no "holding time" within a region (since separate epitaxial regions
would not per se exist). The effect would be to more quickly capture and
sweep upward freed electrons for collection by n-well 430.
The following section will now describe differential QE modulation, and the
advantages that it can provide. Again, QE modulation including
differential QE modulation may be practiced using convention MOS-type
photodiode detectors and/or photogate detectors.
Referring again to FIGS. 5A and 5B, assume that incoming photon energy
generates electron-hole pairs within the substrate of the photodiode
shown, including an electron-hole pair EH.sub.1 generated at an arbitrary
location "X". In FIG. 5A, location X is in the quasi-neutral region and
not in the depletion region (shown cross-hatched). In the present
invention, it is desired that modulation reduce QE at this point in time
and discard as many electron-hole pairs as possible, including EH.sub.1.
If the photodiode QE is then immediately increased, e.g. by increasing
photodiode reverse bias, the depletion region width W can increase to
encompass location X (see FIG. 5B).
In FIG. 5B, EH.sub.1 is still lingering at location X, which is now in the
depletion region, and EH.sub.1 will now contribute strongly to the
photocurrent. On one hand, the increased depletion region in FIG. 5B can
enhance photon detection sensitivity. But electron-hole pairs generated
when photons arrive when QE should be low (FIG. 5A) can contribute to the
total photodiode current when QE should be high (FIG. 5B), e.g., the
contribution is at a different point in time. The undesired result is an
inability to change the effective QE at high modulation rates. But what is
desired is that only photons arriving at a high QE time should contribute
to the photocurrent at any time.
It is desirable to achieve faster photodiode QE modulation by removing the
above-described time lag effect. It is further desirable to remove common
mode effects in the photodiode output signal resulting from ambient light
and from so-called photodiode dark current. Overall, it will now be
appreciated that QE modulation essentially modulates the size of the
collection target for electrons within the photodiode structure. Absent
another collection target, most electrons would eventually be collected by
even a small target due to their relatively long lifetime. Thus, QE
modulation in terms of change in numbers of electrons will be
substantially smaller than the change in target area.
Various aspects of the present invention will now be described that use
differential QE modulation techniques in which the collection target size
is increased and decreased, while the alternative adjacent target size is
decreased and increased. The effect is to provide a larger alternate
target to electrons or holes, while reducing the target area of the given
photodiode. This enhances QE as the electrons will be collected by the
alternative target and taken out of circulation for the reduced target,
well before the end of their lifetime.
During QE modulation, the present invention recognizes that some regions
within a photodiode, typically within the more lightly doped region of the
junction, alternate between quasi-neutral and depletion regions. If these
regions can be kept to a minimum, the photodiode can be more sharply QE
modulated. Such enhanced QE modulation is promoted using a differential
modulation approach, as will be described later herein with respect to
FIGS. 8A and 8B. FIGS. 8A and 8B represent "snapshots" in time of two
adjacent photodiodes, denoted A and B, 180.degree. apart. Preferably
within array 230, adjacent photodiodes A and B are sufficiently close
together and small in surface area such that each receives substantially
the same amount of incoming photon energy at any given time. Photodiodes
groups or banks A and B are bias-modulated such that their respective QE
are 180.degree. out of phase, i.e., QE of photodiode A reaches a maximum
when the QE of photodiode B is at a minimum, and vice versa.
Note in FIGS. 8A and 8B that the quasi-neutral region 500 between adjacent
photodiodes A and B is always quite small, and hence the number of
electron-hole pairs created therein will be quite small. This is
advantageous since it is the quasi-neutral region near the depletion
region that reduces QE modulation. In FIG. 8B, electron-hole pairs in
quasi-neutral region 500 between diodes photodiodes A and B may be swept
into the photocurrent for adjacent photodiode B when QE for photodiode B
is increased. Because quasi-neutral region 500 is small, degradation of QE
modulation due to region 500 will advantageously be small.
Assume in FIGS. 8A and 8B that at a given time photodiodes A and B are
reverse biased at 0 VDC and 2 VDC, respectively. As an example, if A and B
are fabricated with reasonable CMOS 0.25 .mu.m processes, photodiode B
typically will measurably convert up to 30% more photon energy than
photodiode A. The QE of photodiode A goes up rapidly from 0 VDC with small
increases in reverse bias, whereas the QE of photodiode B reverse biased
at say 1 VDC will be almost unaffected by a small change in reverse bias.
Thus, it is advantageous for maximum QE modulation that reverse bias of
photodiode A be as low as possible. This bias regime corresponds to a MOS
transistor whose channel is formed in the quasi-neutral region 500 between
photodiodes A and B. The MOS transistor gate structure is non-existent but
may be assumed to be present at some voltage in sub-threshold regions with
a high source-drain voltage.
During the time frame shown in FIG. 8A, photodiode A is weakly reverse
biased. As a result, substantial leakage current can exist between
photodiodes A and B, which would correspond to sub-threshold leakage of a
MOS transistor whose source is photodiode A and whose drain is photodiode
B in FIGS. 8A and 8B. Such leakage current may be reduced by forming a
polysilicon gate G', assumed transparent to optical energy of interest, at
least over the region between photodiodes A and B, with an insulating
layer of thin oxide (TOX) beneath gate G'. If such a gate is fabricated,
sub-threshold leakage current can be controlled by controlling the gate
voltage. For example, each 0.1 mV of gate voltage corresponds to a
ten-fold change in leakage current. For an undoped channel, a gate voltage
of about -0.4 VDC is typically sufficient to substantially reduce leakage
current.
FIG. 8C is a top view of a portion of array 230 depicting rows and columns
of photodiodes, here labeled as either photodiodes A or photodiodes B. As
suggested by the different cross-hatching, QE modulation nodes for all
photodiodes A are coupled together in parallel, and QE modulation nodes
for all photodiodes B are coupled together in parallel. Essentially, FIG.
8C may be seen as a top view of one large photodiode A and one large
photodiode B. In a differential QE mode of the present invention, all
photodiodes A can be modulated with a phase 180.degree. from the signal
that modulates all photodiodes B. Both classes of photodiodes, e.g. A and
B, will have their respective QE sharply modulated because only a very
small quasi-neutral region will exist between them. It is substantially
only the quasi-neutral region at the bottom region of each photodiode that
causes significant smearing of the QE modulation at high modulation
frequencies.
Having presented an overview of concepts underlying QE modulation, various
configurations of systems employing such techniques will now be described.
In a first category of embodiments, the present invention uses variable
phase delay (VPD) techniques in which dedicated electronic mixers (e.g.,
Gilbert cells) mixers are replaced by QE modulation. System topography
depicting the first category is found primarily in FIGS. 9A-9C. A second
category provides embodiments that mix with fixed phase delays using QE
modulation, and implements a variety of spatial and temporal multiplexing
approaches. System topography depicting the second category is found
primarily in FIG. 10.
Advantageously, either category of embodiments can modulate QE of
MOS-implemented photodiodes by changing photodiode reverse bias, or by
providing MOS-implemented photodiodes with a photogate, and then changing
the gate voltage. Single-ended or double-ended differential signal
processing may be employed with both methods. Differential QE modulation
advantageously allows faster QE modulation, and provides a differential
output that substantially removes common mode effects due to ambient light
and photodiode dark current. Both categories can advantageously accumulate
photodetector signal charge on a photodiode capacitor. Each category can
examine charge periodically when QE modulation is stopped. Such signal
accumulation approaches are preferred over methods that seek to directly
measure a high frequency small magnitude photocurrent.
FIGS. 9A-9C will now be described with respect to various variable phase
delay (VPD) QE modulation embodiments of the present invention, so-called
category one embodiments. Using VPD techniques, photocurrent from each
QE-modulated pixel photodiode (or photogate photodiode) is coupled as
input to an associated relatively high input impedance amplifier that need
not exhibit high bandwidth, high frequency response, or high closed-loop
gain. The amplifier output feeds directly to a low pass filter (LPF) whose
output drives an integrator. The integrator output is coupled as to
control phase of the variable phase delay (VPD) that controls QE
modulation signals that drive the photodetector diodes. The VPD is also
driven by a signal from the periodic signal generator that controls the
optical energy emitter. There may or may not be a DC offset associated
with the output signal from the pixel photodiode detectors and with the
homodyne drive signal. Assuming no offsets, at steady-state the LPF output
will be zero. Assuming appropriate DC offsets, at steady-state the LPF
output will be a minima or a maxima. This method may be implemented
single-ended, or preferably double-ended using a complementary approach in
which positive and negative signals are derived from photodiodes that are
QE modulated out of phase.
For ease of illustration, explicit biasing of photodiode (or photogate)
detectors is not shown. Those skilled in the art will recognize that
providing biasing may be as simple as coupling a resistor from a reference
source to a node on the various photodetectors for single-ended and for
differential mode QE modulation. More preferably, in the case of
differential QE modulation, feedback would be provided to a common mode
biasing reference to ensure that the sum of the two signals being compared
remains within a desired dynamic range.
Referring now to FIG. 9A, a category one variable phase delay (VPD)
embodiment will be described. FIG. 9A depicts a portion of IC 210, array
230, pixel detectors 240-1 through 240-x, and each diode's associated
exemplary electronics 250'-1 through 250'-x. Elements in FIG. 9A that bear
like reference numerals to elements in earlier figures herein may, but
need not be, identical. For example, variable phase delay unit 320 or
filter 340 in FIG. 9A may, but need not, be identical to the same
components in FIG. 4. Each pixel diode 250-x in FIG. 9A has an associated
electronic circuit, denoted 250-x (as contrasted with the notation 250'-x
for FIG. 4). Again for ease of illustration, only two out of perhaps many
thousands of pixel diodes 240 and associated electronic circuits 250 are
depicted. Again, if desired a dedicated A/D converter can be provided as
part of each electronics circuit 250-1 through 250-x, as opposed to
implementing an omnibus A/D function on IC chip 210.
Comparing the configuration of FIG. 4 with that shown in FIG. 9A, it is
seen that whereas FIG. 4 provided each pixel diode with a dedicated
electronic mixer 310, no such separate or explicit mixers are included in
electronics 250-x in FIG. 9A. Instead, according to the present invention,
the configuration of FIG. 9A uses QE modulation to derive phase difference
between transmitted and received signals, and to derive TOF, among other
data. FIG. 9A and other QE modulation embodiments described herein
advantageously avoid mixers and their need for a sufficiently amplified
signal to be input for mixing.
In FIG. 9A, the detected waveform signal photodiodes 240-x in array 230
will include a DC-offset of the form
1+A.multidot.cos(.omega..multidot.t+.phi.), such as shown in FIG. 2C. The
1+A.multidot.cos(.omega..multidot.t+.phi.) signal will preferably have a
minimum value of 0 VDC and a maximum value of perhaps +3 VDC. As noted
earlier with respect to FIG. 2C, the change of notation to include an
arbitrary DC-offset will not impact the relevant mathematical analysis.
In FIG. 9A. the output signal from variable phase delay (VPD) 320 is
coupled via capacitor C.sub.O to node N.sub.d of the associated photodiode
240-x, for each electronics system 250-x in array 230. When C.sub.0
-coupled modulation signal is in phase with the detected light energy,
e.g., S.sub.2 =A.multidot.cos(.omega.t+.phi.), the signal developed across
amplifier 400's input impedance R.sub.i will be maximum. R.sub.i is large,
e.g., >1 G.OMEGA., and the signal voltage across R.sub.i will build-up
in magnitude slowly over a large number of cycles of the periodic signal
cos(.omega.t). The feedback path within each electronics 250-x includes
low pass filter 340 and integrator 330, and the resultant feedback seeks
to minimize magnitude of amplifier 400 input, e.g., the voltage across
R.sub.i. Minimal amplitude across R.sub.i occurs when signal S.sub.2
=A.multidot.cos(.omega.t+.phi.) received by the photodiode 240-x is
180.degree. degrees out of phase with the modulating signal
cos(.omega.t+.psi.). As shown in FIG. 5, for each electronics 250-x, a
resultant phase value .psi..sub.x can be read-out as a voltage signal at
the output of each integrator 330.
Thus electronics 250-x in FIG. 9A functions somewhat similarly to
electronics 250'-x in FIG. 4 to examine incoming periodic photon energy
signals, and to produce a phase output signal from which distance z from
the system to a target object 20 may be measured. In FIG. 9A, each
amplifier output is passed directly to the input of low pass filter 340,
and thus a high frequency response for amplifiers 400 is unnecessary.
Moreover, the voltage signal across each amplifier input impedance R.sub.i
is allowed to build-up over a large number of periodic cycles. Thus, the
final signal to be detected will be relatively large, e.g., preferably
many mV or tens of mV. As a result, unlike amplifiers 300 in FIG. 4, in
the embodiment of FIG. 9A, amplifiers 400 need not be very high gain, very
low noise, high frequency devices. As a result, amplifiers 400 can be
implemented in less IC chip area and will consume less current, yet can
help provide better z-distance resolution than the more complicated
configuration of FIG. 4.
Turning now to FIG. 9B, an additional category one VPD embodiment is
depicted. In FIG. 9B, complementary, 180.degree. out of phase, outputs
from VPD 320 are employed, in which one VPD output is coupled via a
capacitor C.sub.O to an associated photodiode D or 240-x. The
complementary VPD output is coupled via a similar capacitor C.sub.10 to a
similar photodiode, here denoted D'. Thus, photodiode 240-x is QE
modulated by one VPD output, whereas diode D' is QE modulated 180.degree.
out of phase by the other VPD output. In essence, QE modulation nodes for
various photodiodes are parallel-coupled such that groups of photodiodes
are parallel QE modulated. Photodiodes 240-x and D' each discharge, and
there will be a common mode signal requiring that reverse bias voltages to
each photodiode be refreshed periodically to a predetermined level.
Further, the configuration of FIG. 9B uses differential inputs to
amplifiers 400', the effects of ambient light falling upon photodiodes
240-x in array 230 are minimal. An additional advantage provided by the
configuration of FIG. 9B is that photodiodes 240-x and associated
photodiodes D' can be implemented with a differential structure that
enables rapidly modulating QE for the diodes sets without significant lag.
Thus, for each photodiode 240-x in array 230, a photodiode D' having
substantially identically characteristics will be coupled to the inverting
input (in the configuration of FIG. 9B) of each amplifier 400'.
Turning now to FIG. 9C, a VPD QE modulation embodiment employing
differential comparators and digital integrators is shown. Again it is
understood that QE modulation nodes for various photodiodes are
parallel-coupled such that photodiodes can be parallel QE modulated. In
FIG. 9C, amplifiers 400' and typically analog integrators 330 of FIG. 9B
are replaced with differential comparators 510, and with digital
integrators 520. At regular intervals, microcontroller 260 (see FIG. 3)
will command energy emitter 220 to halt emission, or to shut down, and
both outputs of VPD 320 will be set to a constant voltage. Each
differential comparator 510 then compares the differential signals
presented to its input nodes. Each digital integrator 520 then reads the
result (C) of this comparison, and increments its digital output by a
small amount if C=1 and reduces its output by a small amount if C=0. If
desired, comparators 510 can be shut down when the photodiodes are being
modulated, during which times voltage comparisons are not required.
Referring still to FIG. 9C, consider the following example. At
steady-state, the output signal from digital comparator 510 will toggle
between "0" and "1". The output from digital integrator 520 will continue
toggling between two values, e.g., 5 and 6. VPD unit 320 will produce
delays, toggling between 5 and 6 (in the present example). Photodetector D
will continue to be modulated with a signal that toggles between
cos(.omega.t+5) and cos(.omega.t+6). In the above example, if values 5 and
6 are sufficiently close, at equilibrium it will appear as though
photodiode D were being modulated by cos(.omega.t+5.5).
So-called category two embodiments employing fixed phase QE modulation will
now be described primarily with reference to FIG. 10. In category two
embodiments, fixed phase signals are used to QE modulate each
photodetector. Different groups or banks of photodiode detectors may be
defined in a non-localized manner within the array. For example, a first
bank of photodiode detectors may be QE modulated with fixed 0.degree.
phase shift, a second bank may be QE modulated with fixed 90.degree.
phase, shift, a third bank with fixed 180.degree. phase shift, and a
fourth bank with fixed 270.degree. phase shift. Within each pixel, there
are photodiode detectors that correspond to every one of the four banks.
Phase information and target object brightness information can be
determined by examining output values for each bank within a pixel. This
fixed delay approach simplifies the electronic circuitry associated with
each pixel, reduces power consumption, can reduce IC chip area
requirement, and enables a range of techniques for temporal and spatial
multiplexing.
Various aspects of category two QE modulation will be described including
spatial and temporal multiplexing, which multiplexing may be single-ended
or differential, as well as none one-to-one mapping between physical
photodetectors and pixels. Further, category two embodiments can employ an
inductor to reduce power consumption by tuning-out or compensating for
capacitive losses.
Category two fixed phase delay QE modulation will now be described with
reference to FIG. 10. An advantage of this configuration is that
electronics 250-x can be somewhat simplified and, as in other QE
modulation embodiments, a brightness measurement can be output. In FIG.
10, photodiodes 240-x and D' in array 230 are modulated with a fixed phase
modulator 530 whose output is selectable, e.g., by microcontroller 260
(see FIG. 3) to be 0.degree. phase or 90.degree. phase. Software that may
be included within memory 270 preferably corrects for the (fixed)
modulation phase differences between pixel photodiodes due to path delays
to the pixels. The modulating signal and its complement may be provided to
pixel array 230, or the complement may be regenerated within each pixel
electronics 250-x by including a 180.degree. delay unit 540 coupled to the
single output of a fixed phase delay unit 530.
In FIG. 10, system 200 (see FIG. 3) is permitted to operate for a large
number of cycles (where core frequency is .omega.), after which the laser
or other photon energy emitter 220 is shut down. When emitter 220 is shut
down, the diode modulating voltage signal and its complementary signal are
set to a fixed magnitude. In the following description, so-called
"cos(.omega.t)+1" analysis will be used. Assuming that QE modulation is
somewhat linear, the result of multiplying the photodiode (D) signal
(B{cos(.omega.t+.phi.)+1}) with the modulating signal (cos(.omega.t)+1))
and then integrating is B(0.5{cos(.phi.)}+1). The result of multiplying
the photodiode (D') signal (B{cos(.omega.t+.phi.)+1}) with the modulating
signal (cos(.omega.t+180.degree.)+1)) is B(-0.5{cos(.phi.)}+1).
Subtracting the two expressions will then yield at the output of
differential amplifier 400' the signal V.sub.0 =B.multidot.cos(.phi.),
where B is a brightness coefficient. A new measurement is then carried out
with the modulation phase 90.degree. apart from the original modulating
signal. The result at the output of amplifier 400' will then be V.sub.90
=B.multidot.sin(.psi.). From the 0.degree. and 90.degree. measurements,
angle .psi. can be obtained from:
tan(.psi.)=V.sub.90 /V.sub.0.
The brightness B can be obtained from
B=V.sub.0.sup.2 +V.sub.90.sup.2
Advantageously, and in contrast to the embodiments described earlier
herein, the configuration of FIG. 10 does not require an integrator within
each electronics 240-x, thereby simplifying the system design.
A further advantage of the configuration of FIG. 10 is that
impedance-matching inductors may be employed to reduce system operating
power. For example, assume each photodiode 240-x is about 15 .mu.m square
and has capacitance (C) of about 10 FF. Assume too that the modulating
frequency f, where f=.omega./(2.PI.), is about 1 GHz, and that system 200
is operated from a 3 VDC power source (V), for example a battery supply.
Power consumption per photodiode pixel will be proportional to
C.multidot.V.sup.2.multidot.f and will be about 8 .mu.W. For an array 230
comprising 200 pixels .times.200 pixels, power consumption will be about
0.32 W.
Since power consumption is directly proportional to capacitance C, power
consumption can be reduced by decreasing the effective capacitance. This
desired result is achieved by coupling a tuned inductor (L.sub.p) in
parallel with the capacitance of the photodiodes. However if tuned
inductors L.sub.p were placed inside each pixel as shown in FIG. 11A, to
resonant at 1 GHz, each inductor L.sub.p would be on the order of 100
.mu.H, far too large a value to implement within each pixel photodiode.
In contrast to the VPD QE modulation embodiment of FIG. 9C, in the
embodiment of FIG. 10, all pixels are modulated using a common modulation
signal for each parallel-coupled bank of photodiodes, akin to photodiodes
A and B in FIG. 8C. An advantage of this configuration is that all
photodiodes in a bank of parallel-coupled photodiodes be driven in
parallel. The various parasitic shunt capacitances for each
parallel-coupled photodiode are themselves coupled in parallel. The result
is that one (or relatively few) inductors need be parallel-coupled to all
photodiodes in a parallel-bank to achieve resonance at the desired
frequency. In the above example of a 200.times.200 array, 100 .mu.H would
be required for each pixel photodiode. By parallel-coupling say
200.times.200 photodiodes lowers the value of L.sub.p to 100
.mu.H/(200.multidot.200) or 0.25 nH, a very realistic magnitude of
inductance to fabricate. Further, array sizes may indeed be larger than
200.times.200, in which case the overall capacitance of a greater number
of photodiodes increases, which further reduces the magnitude of the
single inductor L.sub.p required to resonate at the desired QE modulation
frequency. Such inductance may be fabricated on IC chip 210 or even
mounted off-chip. For the above example, a single inductor L.sub.p in FIG.
11B on the order of 0.25 nH would tune-out the effective capacitance of
the 200.times.200 photodiodes that are parallel-coupled, whereas in FIG.
11A, each photodiode would require a separate inductor of substantially
greater inductance.
The fixed phase delay (category two) configuration of FIG. 10 is intended
to be exemplary. In practice, various so-called spatial multiplexing and
temporal multiplexing techniques may be employed. Different spatial
topologies (of which differential QE modulation shown in FIG. 8C is but
one example) can be used to refer to different groups or banks of
photodetectors within the array that can be modulated group-wise with a
fixed phase. Spatial topology can enhance collection of photon-energy
released charges within the photodetectors, and thus can enhance signal
detection. Temporal topology refers to modulating the same bank of
photodetectors with different fixed modulation phases at different times.
Some spatial topologies permit spatial multiplexing, which can include the
sharing of photodetectors across multiple pixels, e.g., the re-using of a
same photodetector in different pixels. Temporal topology can give rise to
multiplexing in time, which can promote pipelining. The present invention
can implement any or all of the aspects, with various pixel bank
topologies, and with various time-phase topologies.
The spatial multiplexing technique embodied in FIG. 8D is what is shown in
the exemplary of configuration of FIG. 10, in which the photodetector
topology was that of FIG. 8C, and in which a 0.degree.-180.degree.,
90.degree.-270.degree. time topology was used. Further, the exemplary
configuration of FIG. 10 may also be used to support spatial-multiplexing
of the photodiodes, as well as time-multiplexing or pipelining.
A different spatial topology embodiment of the present invention will now
be described with reference to FIG. 12A. The spatial-multiplexing
embodiment of FIG. 12A operates in principle similarly to the
0.degree.-180.degree.-90.degree.-270.degree. time-division topology
embodiment of FIG. 10. The difference, however, is that measurements are
now obtained simultaneously at time .tau..sub.1, for example using four
photodetectors d.sub.1 or 240-(x), d.sub.2 or 240-(x+1), d.sub.3 or
240-(x+2), and d.sub.4 or 240-(x+3), shown in plan view in FIG. 12A.
As before, .DELTA.V.sub.d =[.DELTA.V.sub.d1 (.tau..sub.1)-.DELTA.V.sub.d2
(.tau..sub.1)]/[.DELTA.V.sub.d3 (.tau..sub.1)-.DELTA.V.sub.d4
(.tau..sub.1)]=tan(.phi.).
Turning now to FIG. 12B, it will be appreciated that photodetectors may be
shared in different pixels across the photodetector array. In FIG. 12B,
the four detectors shown in FIG. 12A are depicted with cross-hatching so
that their dual-role can be seen. For example, photodiodes d1-d2-d3-d4 may
be said to form a cluster of four photodetectors within a pixel in array
230. However, photodiodes d1 and d3 are also members of a photodiode
cluster comprising photodiodes d1, d5, d3, d6, and so on. Note that while
individual photodetectors can play multiple roles in different clusters,
no additional IC chip area is required to implement the
spatially-multiplexed embodiment shown, thus promoting efficient use of IC
chip area. If desired, additional data measurements are obtained by
reusing partial measurements in space.
If desired, a 0.degree.-120.degree.-240.degree. time-division QE modulation
embodiment may be carried out, although this may not be the most efficient
embodiment. In such embodiment two measurements taken from the array of
pixels shown in FIG. 8C at time frames .tau..sub.1 and .tau..sub.2 are
used. For the first measurement at time .tau..sub.1, a photodetector bank
(bank A) comprising photodetectors A is enabled with an S1(t) sinusoidal
waveform at 0.degree. phase, while adjacent photodetector bank (bank B)
comprising photodetectors B is de-phased 120.degree. by S2(t). For the
second measurement at time .tau..sub.2, bank B is de-phased 120.degree.
and bank A is de-phased 240.degree.. The total phase difference is derived
as follows:
.DELTA.V.sub.d =[.DELTA.V.sub.d2 (.tau..sub.2)-.DELTA.V.sub.d1
(.tau..sub.2)]/.DELTA.V.sub.d1 (.tau..sub.1),
where at time .tau..sub.1
.DELTA.V.sub.d1 =A [1+cos(.omega.t)]cos(.omega.t+.phi.)
.DELTA.V.sub.d1 =A
cos(.omega.t+.phi.)+0.5A{cos(.phi.)+cos(2.omega.t+.phi.)}
and at time .tau..sub.2
.DELTA.V.sub.d1 =A[1+cos(.omega.t-120)]cos(.omega.t+.phi.)
.DELTA.V.sub.d1 =A
cos(.omega.t+.phi.)+0.5A[cos(.phi.-120)+cos(2.omega.t+.phi.+120)]
.DELTA.V.sub.d2 =A[1+cos(.omega.t-240)]cos(.omega.t+.phi.)
.DELTA.V.sub.d2 =A cos(.omega.t+.phi.)+0.5A[
cos(.phi.-120)+cos(2.omega.t+.phi.+120) ]
hence, after filtering
.DELTA.V.sub.d =[cos(.phi.-120)-cos(.phi.+120)]/cos(.phi.)
.DELTA.V.sub.d =2 sin(.phi.)sin(120)/cos(.phi.)
.DELTA.V.sub.d =K.sub.1 sin(.phi.)/cos(.phi.)
.DELTA.V.sub.d =K.sub.1 tan(.phi.), where K.sub.1 =3.
Referring now to FIG. 12C, a 0.degree.-120.degree.-240.degree. modulation
(spatial-multiplexing) embodiment is shown. This spatial-multiplex
embodiment is similar to the above-described
0.degree.-120.degree.-240.degree. time-division multiplex embodiment
except that measurements are obtained simultaneously at time .tau..sub.1
using three detectors d.sub.1, d.sub.2, and d.sub.3. As above,
.DELTA.V.sub.d =[.DELTA.V.sub.d3 (.tau..sub.1)-.DELTA.V.sub.d2
(.tau..sub.1)]/.DELTA.V.sub.d1 (.tau..sub.1)=K.sub.1 tan(.phi.), where
K.sub.1 =3.
It will be appreciated from what has been described with respect to FIG.
12B, that photodetectors in FIG. 12C may be shared across different pixels
in photodetector array 230.
Referring back to FIG. 8C, it will be appreciated that each photodetector
in a bank A can be shared across four pixels, e.g., top and bottom, left
and right. For example, in the second row of photodetectors, the first
detector A may be associated with each of the four adjacent detectors B.
It will be appreciated that to facilitate spatial multiplexing according to
the present invention, it can be advantageous to obtain raw data
single-endedly from each photodetector, rather than obtain differential
data, e.g., first generate a difference signal between banks of photodiode
detectors. QE modulation is preferably still carried out differentially,
i.e., with multiple banks of detectors modulated with different phases.
Such single-ended raw data can be preferable in that greater flexibility
can exist in signal processing the data, e.g., adding or subtracting data
from adjacent photodetectors (e.g., perhaps digitally), than if only
differential data were available. FIG. 13A shows typically differential
signal processing of photodetector output whereas FIG. 13B shows
single-ended signal processing.
The concept of pipelining with respect to embodiments such as shown in FIG.
10 will now be described. As used herein pipelining refers to reduction of
latency in obtaining pixel measurements in successive frames of acquired
data.
One can interlace measurements within frames of acquired data to increase
measurement throughput as follows:
0.degree.-180.degree. measurement: .DELTA.V.sub.d (.tau..sub.1)
90.degree.-270.degree. measurement: .DELTA.V.sub.d
(.tau..sub.2).fwdarw..DELTA.V.sub.d (.tau..sub.2)/.DELTA.V.sub.d
(.tau..sub.1)=tan(.phi.)
0.degree.-180.degree. measurement: .DELTA.V.sub.d
(.tau..sub.3).fwdarw..DELTA.V.sub.d (.tau..sub.2)/.DELTA.V.sub.d
(.tau..sub.3)=tan(.phi.)
90.degree.-270.degree. measurement: .DELTA.V.sub.d
(.tau..sub.4).fwdarw..DELTA.V.sub.d (.tau..sub.4)/.DELTA.V.sub.d
(.tau..sub.3)=tan(.phi.), etc.
In this fashion, a continuous pipeline of measurement information can be
computed with an effective doubling of computational speed, yet with a
latency of one measurement. Indeed, one advantage of the above-described
time-division multiplexing QE modulation embodiment is that frame rate of
data acquisition is substantially increased. As noted, on-chip CPU system
260 may be used to perform the information processing steps described
herein, and on-chip electronics 250-x can implement that various forms of
QE modulation and signal processing that have been described.
Referring once again to FIG. 8A, assume that each of the two side-by-side
photodetectors 240-(x) (or detector "A") and 240-(x+1) (or detector "B")
have substantially identical area when seen in a planar view. What will
now be described are techniques for reducing mal-effects of non-uniform
illumination falling upon these photodetectors, including effects
associated with differences in actual photodetector effective areas, and
also reducing 1/f noise associated with gain of the amplifiers used with
these photodetectors.
Referring to FIG. 3 and to FIG. 8A, assume that photon energy returned from
target object 20 falls upon photodetectors A and B, and that these two
photodetectors output different signals, e.g., different magnitudes. The
detected output signal may be different for several reasons. Perhaps the
illumination falling upon photodetector A differed from the illumination
falling upon photodetector B. Perhaps the effective detection area of
photodetector A differed from photodetector B due to component
mismatching, or perhaps photodetector A was simply better fabricated and
exhibits better detection characteristics.
Referring again to the embodiment of FIG. 10, and using a "1+cos" analysis
for simplicity of explanation, let the incoming photon energy signal seen
by photodetector A is A'{cos(.omega.t+.phi.)+1} and let the incoming
photon energy signal seen by detector B be B'{cos(.omega.t+.phi.)+1}. If
A'=B', there is uniform illumination, but not otherwise. The more general
case, however, results where A' and B' are not identical.
In FIG. 10, the energy signal seen by detector A,
A'{cos(.omega.t+.phi.)+1}, is multiplied by {cos(.omega.t)+1} to yield
after accumulation A'(0.5cos(.phi.)+1), hereafter denoted expression {1}.
Similarly, the energy signal seen by detector B,
B'{cos(.omega.t+.phi.)+1}, is multiplied by {cos(.omega.t+180.degree.)+1}
to yield, after accumulation, B'(-0.5cos(.phi.)+1), hereafter denoted
expression {2}. If A'=B', then it would be a simple matter to obtain A'
cos(.phi.), as described earlier herein. The problem is that A' and B' are
not equal.
In the earlier description of FIG. 10, a goal was to arrive at
Kb{cos(.phi.)} and Kb{sin(.phi.)}, where Kb is a brightness coefficient.
For the case of non-uniform illumination, the present invention now
multiplies A'(cos(.omega.t+.phi.)+1) by {cos(.omega.t+180.degree.)+1},
which after integration yields A'(-0.5cos(.phi.)+1), hereafter expression
{3}. Further, the present invention also multiplies
B'{cos(.omega.t+.phi.)+1} by {cos(.omega.t)+1} to yield B'(0.5
cos(.phi.)+1), hereafter expression {4}.
At this juncture, the present invention performs the mathematics to carry
out (expression {1}-expression {2})-(expression {3}-expression {4}), to
arrive at (A'+B'){cos(.phi.)}. Similarly the same operation can be carried
out to arrive at the equivalent (A'+B'){sin(.phi.)}, as noted earlier with
respect to FIG. 10.
Thus, one calculation may be carried out upon (expression {1}-expression
{2}) and a similar calculation carried out upon (expression {3}-expression
{4}). Schematically, the procedure may be carried out as follows,
referring now to FIG. 8A, FIG. 10, and FIGS. 14A and 14B:
(1) at time 0<t<t1, detector D or 240-(x) is biased with signal
S1=1+cos(.omega.t) and detector 240-(x+1) is biased with signal
S2=1+cos(.omega.t+180.degree.), e.g., 0.degree. and 180.degree.
modulation;
(2) signals output from the two detectors are accumulated during time
0<t<t1 and at time t=t1, the differential signal is stored or
sampled in digital or analog form;
(3) during time t1<t<t2, detector 240-(x) is biased with signal
S1=1+cos(.omega.t+180.degree.) and detector 240-(x+1) is biased with
signal S2=1+cos(.omega.t);
(4) output signals from the two detectors are accumulated, and at the end
of accumulation at time t=t2, the differential signal is stored or
sampled, in digital or analog form; and
(5) a difference signal is computed for the analog and/or digital signals
that have been sampled or stored.
FIGS. 14A and 14B depict exemplary techniques for signal subtraction in the
analog domain and in the digital domain, respectively. The analog or
digital "shared" components 700 may be placed outside the photodiode pixel
detector, perhaps using one shared component per each column in the
row-column array of pixel detectors. Sample and hold (S/H) units within
the pixel will hold both measurements for the entire duration of a
read-out operation, which operation is repeated independently for each row
of pixels. Alternatively, one might perform averaging and even the
analog-to-digital (ADC) conversion within the pixel block.
In FIG. 14A, the shared circuitry 700 includes an analog summer 710 whose
analog output is digitized by an analog-to-digital converter 720. In FIG.
14B, the shared circuitry is essentially a digital adder 730 whose inputs
are negated. The output from adder 730 is input to a register 740 whose
output is fedback to an input of the adder. An A/D converter 720 presents
digital input to the adder. In FIG. 14B, averaging is carried out in the
digital domain, and analog-digital conversion can be shared across all
rows of pixels, which means a S/H will be required per pixel to hold the
accumulated voltage signal before the signal is delivered to the ADC for
conversation. Thus in the digital domain embodiment of FIG. 14B, signal
averaging requires twice as many A/D conversions than in the analog domain
embodiment of FIG. 14A. It will be appreciated that similar approaches can
be used in the various other modulation schemes that have been described,
including time-division multiplexing, and spatial multiplexing.
In the various embodiments described herein, movement of objects within a
detected image contour can be computed, e.g., by microprocessor 260, by
identifying contour movements between frames of acquired data. The pixel
detectors within the contour can all receive a uniform velocity that is
the velocity of the contour. Since objects can be identified using their
contours, one can track objects of interest using the on-chip processor
260. As such, if desired IC chip 210 can export a single value (DATA) that
can represent change in location of the entire object 20 whenever it has
moved. Thus instead of exporting from the IC chip an entire frame of
pixels at the frame rate, a single vector representing the change in
location of the object of interest may instead be sent. So doing results
in a substantial reduction in IC chip input/output and can greatly reduce
off-chip data processing requirements. It will be appreciated that the
on-chip microprocessor 260 can also supervise sequencing of spatial and/or
temporal topologies, and can also optimize spatial and/or temporal
multiplexing.
In other applications, system 200 may be called upon to recognize an object
that is a virtual input device, for example a keyboard whose virtual keys
are "pressed" by a user's fingers. For example, in co-pending U.S.
application Ser. No. 09/502,499, filed Feb. 11, 2000, and entitled "Method
and Apparatus for Entering Data Using a Virtual Input Device" a
three-dimensional range-finding TOF system is used to implement virtual
input devices. As a user's hand or stylus "presses" a virtual key or
region on such device, the system using TOF measurements can determine
which key or region is being "pressed". The system can then output the
equivalent of key stroke information to a companion device, for example a
PDA that is to receive input data from the interaction of a user with the
virtual input device. The present invention may be used in such
application, in which case DATA in FIG. 3 could represent keystroke
identification information that has been processed on-chip by
microprocessor 260.
As noted, microprocessor 260 executing software perhaps associated with
memory 270 can control modulation of generator 225 and detection by the
various electronic circuits 250. If desired, detection signals may be
processed using special image processing software. Since system 200
preferably can be battery operated due to its low power consumption, when
such software determines that sufficient image resolution is attained,
operating power may be terminated selectively to various portions of array
230. Further if sufficient photon energy reaches array 230 to ensure
adequate detection, the shape of signals output by emitter 220 could be
changed. For example, the peak power and/or duty cycle of the emitter
energy could be reduced, thus reducing overall power consumption by system
200. The design tradeoffs in changing the shape of the optical energy
output signal involve considerations of z-resolution accuracy, user
safety, and power handling capacity of emitter 220.
In summary, the overall system advantageously can be operated from a small
battery in that peak and average power from optical emitter 220 is
preferably in the tens of mW range. Nonetheless distance resolution is in
the cm range, and signal/noise ratios are acceptable. Although various
embodiments have been described with respect to acquiring information
proportional to distance z, it will be appreciated that, if desired, the
present invention could be practiced to acquire information relating
solely to brightness of a target object. In such an application, the
present invention can be used essentially as a rather good filter that
substantially reduces ambient light effects upon brightness information.
Whereas acquiring z-information may involve modulating an energy source at
a modulation frequency in excess of 100 MHz, an application directed to
acquiring brightness information could modulating the energy source at a
substantially lower rate, perhaps 50 Khz or so.
Modifications and variations may be made to the disclosed embodiments
without departing from the subject and spirit of the invention as defined
by the following claims.
* * * * *